Method and apparatus for processing a television signal with a coarsely positioned if frequency

ABSTRACT

Various embodiments are described herein for a universal television receiver that is capable of processing television channel signals broadcast according to a variety of analog and digital broadcast standards. Analog processing includes using coarse filtering with pass bands that are wide enough to accommodate frequency shifts in a desired television channel signal and analog circuitry variability and digital processing includes tracking a carrier frequency of the desired television channel signal to generate and apply a frequency shift feedback signal to compensate for frequency shifts in the carrier frequency.

REFERENCE TO RELATED APPLICATION

This application claims priority from U.S. Provisional PatentApplication Ser. No. 60/894,832 filed on Mar. 14, 2007 and from U.S.Non-Provisional patent application Ser. No. 12/038,781 filed on Feb. 27,2008.

FIELD

Various embodiments of systems, system blocks and corresponding methodsare described herein that relate to a universal television receiver thatcan process television signals that are broadcast according to differenttelevision broadcast standards.

BACKGROUND

Television signals are broadcast according to several different types oftelevision broadcast standards. These television broadcast standardsinclude variations of NTSC, SECAM and PAL for analog signals, and ATSC,DVB-T and ISDB-T for digital signals. These television broadcaststandards have different characteristics such as bandwidth, modulationtype and the location of audio in the case of analog signals.Accordingly, traditional television receivers have been specificallybuilt to process certain television signals based on a particulartelevision broadcast standard. Traditional television receiverstypically use a SAW filter that has a very sharp frequency response witha passband that corresponds to the bandwidth of the television signalthat is being received. The SAW filter is a relatively expensivecomponent that cannot be integrated onto a chip, and does not readilyallow a television receiver to receive television signals that aretransmitted according to different television broadcast standards,without further specialized processing.

SUMMARY

In one aspect of the invention, at least one of the embodimentsdescribed herein provides a television receiver for processing receivedtelevision signals to provide video and audio information for a desiredtelevision channel signal. The television receiver comprises an analogprocessing block for providing coarse filtering and amplification to amulti-channel television signal to produce a first signal, the coarsefiltering being configured to use pass bands that are wide enough toaccommodate frequency shifts in the desired television channel signaland analog circuitry variability; an analog to digital converter coupledto the analog processing stage for digitizing the first signal toproduce a second signal; and a digital processing block coupled to theanalog to digital converter for processing the second signal to obtainthe video and audio information for the desired television channelsignal. The receiver is configured to track a carrier frequency of thedesired television channel signal and generate and apply a frequencyshift feedback signal to compensate for frequency shifts in the carrierfrequency.

In another aspect of the invention, at least one of the embodimentsdescribed herein provides a method of processing received televisionsignals in a television receiver to provide video and audio informationfor a desired television channel signal. The method comprises:

-   -   providing coarse filtering and amplification to a multi-channel        television signal to produce a first signal, the coarse        filtering being configured to use pass bands that are wide        enough to accommodate frequency shifts in the desired television        channel signal and analog circuitry variability in the        television receiver;    -   digitizing the first signal to produce a second signal, and    -   processing the second signal to obtain the video and audio        information for the desired television channel signal by        tracking a carrier frequency of the desired television channel        signal and generating and applying a frequency shift feedback        signal to compensate for frequency shifts in the carrier        frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

For a better understanding of the various embodiments described hereinand to show more clearly how they may be carried into effect, referencewill now be made, by way of example only, to the accompanying drawingsin which:

FIG. 1A is a spectral diagram of the entire television band;

FIG. 1B is a diagram of a blocking profile showing an exemplary range ofsignal strength for received television signals;

FIG. 2 is a high-level block diagram of an exemplary embodiment of auniversal television receiver;

FIG. 3 is a block diagram of an exemplary embodiment of an RF processingblock that can be used in the universal television receiver;

FIG. 4 is a block diagram of an exemplary embodiment of an analogprocessing block that can be used in the universal television receiver;

FIGS. 5A-5C are spectral plots of signals at various locations in theanalog processing block of FIG. 4;

FIGS. 6A-6B are spectral plots of exemplary filter transfer functionsthat can be used for filters in the analog processing block;

FIG. 7 is a block diagram of an exemplary embodiment of a digitalprocessing block that can be used in the universal television receiver;

FIG. 8 is a flow chart diagram of an exemplary embodiment of a standardsdetection method for detecting television transmission broadcaststandards;

FIG. 9 is a block diagram of an exemplary embodiment of an inputfiltering block that can be used in the digital processing block;

FIG. 10 is a block diagram of an exemplary embodiment of a videoprocessing block that can be used in the digital processing block;

FIGS. 11A, 11B and 11C are spectral diagrams of a general desiredtelevision channel signal, a desired television channel signal accordingto an analog television broadcast standard and a desired televisionchannel signal according to a digital television broadcast standardrespectively;

FIGS. 11D, 11E and 11F show spectral plots illustrating the operation ofa video pre-polyphase filter, a video polyphase filter and a main videofilter that are used in the video processing block;

FIGS. 11G, 11H and 11I show the magnitude, real and imaginary parts of asignal that is processed by a VSB filter of the video processing block;

FIGS. 11J, 11K and 11L show the magnitude, real and imaginary parts ofthe frequency response of the VSB filter of the video processing block;

FIG. 12A is a block diagram of an exemplary embodiment of a carrierrecovery block that can be used in the video processing block;

FIG. 12B is a diagram illustrating the phenomenon of overmodulation;

FIG. 12C is a diagram illustrating a first technique for dealing withovermodulation;

FIG. 12D is a diagram illustrating a second technique for dealing withovermodulation;

FIG. 13A is a block diagram of an exemplary embodiment of an audiofiltering block that can be used in the digital processing block;

FIG. 13B is a block diagram of an alternative exemplary embodiment of anaudio filtering block that can be used in the digital processing block;

FIG. 14 is a block diagram of an exemplary embodiment of an audioprocessing block that can be used in the digital processing block;

FIG. 15 is a flow chart diagram of an exemplary embodiment of a gaincontrol method that may be employed by the universal television receiverof FIG. 3;

FIG. 16A is a block diagram of an exemplary embodiment of the analoggain control block of FIG. 9, which can be used to employ an alternategain control method;

FIG. 16B is a block diagram of a leaky peak detector shown in FIG. 16A;

FIG. 16C is an illustration showing RF/IF take-over for the gain controlmethod used by the analog gain control block of FIG. 16A;

FIG. 17A is a block diagram of another exemplary embodiment of auniversal television receiver;

FIG. 17B is a block diagram of an exemplary embodiment of a digitaltelevision demodulator;

FIG. 18 is a block diagram of another exemplary embodiment of auniversal television receiver;

FIG. 19A is a block diagram of another exemplary embodiment of auniversal television receiver that employs aliasing avoidance;

FIG. 19B is a block diagram of an exemplary embodiment for the variablePhase Lock Loop of FIG. 19A;

FIG. 19C is a spectral plot illustrating interference of a desiredtelevision channel due to aliasing;

FIG. 19D is a spectral plot illustrating the avoidance of an aliasedinterferer onto the desired television channel by using sampling rateadjustment;

FIG. 20A is a spectral plot illustrating interference of a desiredtelevision channel due to mixing distortion products;

FIG. 20B is a spectral plot illustrating the avoidance of distortioninterference of the desired television channel by using a localoscillator frequency shift;

FIG. 21A is another spectral plot illustrating interference of a desiredtelevision channel due to mixing of distortion products; and

FIG. 21B is a spectral plot illustrating the avoidance of distortioninterference of the desired television channel by using a localoscillator frequency shift and sampling rate adjustment.

DETAILED DESCRIPTION

It will be appreciated that for simplicity and clarity of illustration,elements shown in the figures have not necessarily been drawn to scale.Further, where considered appropriate, reference numerals may berepeated among the figures to indicate corresponding or analogouselements. In addition, numerous specific details are set forth in orderto provide an adequate understanding for practicing the variousembodiments described herein. However, it will be understood by those ofordinary skill in the art that the various embodiments described hereinmay be practiced without these specific details. In other instances,some methods, procedures and components have not been described indetail since they are well known to those skilled in the art.Furthermore, it should be understood that the word “exemplary” is usedherein to denote an example embodiment of a device or method and notnecessarily indicate a preferred implementation of a device or method.

Referring now to FIG. 1A, shown therein are the spectral characteristicsof a wideband television signal 10. The wideband television signal 10 isessentially continuous with a positive component ranging from 42 to 862MHz and a corresponding negative component ranging from −42 to −862 MHz.An individual television signal has a bandwidth in the range of 6-8 MHzdepending on the television broadcast standard with which the televisionsignal corresponds. For instance, NTSC television signals that are usedin North America have a bandwidth of 6 MHz, while television signalsthat are used elsewhere may have a bandwidth of 7 or 8 MHz. In somecountries, different bandwidths can be used in different parts of thetelevision band. Other parameters will also vary for differenttelevision broadcast standards as is commonly known to those skilled inthe art.

Television signal quality can be dictated by differences in the power ofthe television signals that are received at the television receiver. Thedifference in power depends on the local geography and the location ofthe transmitters with respect to the television receiver. A measure ofgood television signal quality can be specified in terms of SNR whichdepends on the television broadcast standard. For instance, analog NTSCtelevision signals may benefit from greater than 45 dB SNR for goodsignal quality for a Composite Video Baseband Signal (CVBS) output.Meanwhile, digital television signals may require as little as 15 dBcarrier-to-noise ratio for good signal quality in the case of ATSC,since processing of digital television signals uses error detection andcorrection.

Referring now to FIG. 1B, shown therein is a television signal blockingprofile 20, based on the US ATSC A/74 Receiver Performance Guidelines,which is expected to be a widely accepted standard. FIG. 1B shows thatthere can be a wide range in terms of the strength of a desiredtelevision channel signal 22, relative to the strength of other channelsat nearby frequencies. For instance, according to the ATSC performanceguidelines for digital receivers, the weakest signal strength for thedesired television channel signal 22 that should be correctly receivedis −83 dBm, while the strongest signal strength is −4 dBm. Furthermore,the desired television channel signal 22 may have strong undesiredtelevision channel signals 24 and 26 directly adjacent with a relativepower of 33 dB. Subsequent adjacent television channels 28 to 46 canincrease in power by 4 dB for each additional channel separation until57 dB is reached, at which point reception should still be possible.Conventional television receivers deal with this technical challenge inpart by filtering a major portion of the received television signal byusing a tracking filter right at the input of the tuning stage of thetelevision receiver.

Referring now to FIG. 2, shown therein is a high-level block diagram ofan exemplary embodiment of a universal television receiver 100 that canreceive and process digital and analog television signals that aretransmitted according to a variety of broadcast signals, and thereforehave different television channel signal bandwidths. The televisionreceiver 100 comprises an RF processing block 102, an analog processingblock 104, an analog to digital converter (ADC) 106, a digitalprocessing block 108, and a digital to analog converter (DAC) block 110.The television receiver 100 receives the wideband television signal 10and provides a processed version of the desired television channelsignal 112. The digital processing block 108 provides control signals tothe RF and analog processing blocks 102 and 104 and the ADC 106 as isdescribed in further detail below. Furthermore, depending on thetelevision broadcast standard for the desired television channel signal,the digital processing block 108 can output a modulated digital signal112′. The DAC block 110 can contain several digital to analog convertersdepending on the type of output that is required. For instance, therecan be one DAC for a CVBS output and there can be at least one more DACfor a sound IF output. Alternatively, in other embodiments, some outputsignals provided by the universal television receiver 100 may beprovided to a direct digital connection on a downstream component inwhich case there is no need for a DAC for these output signals.

The RF processing block 102, analog processing block 104 and digitalprocessing block 108 are custom blocks. However, in an alternativeembodiment, an off-the-shelf RF processing block can be used withcorresponding changes in some operating parameters and processing in theanalog and digital processing blocks 104 and 108 and the ADC 106. Thisalternative embodiment is described in further detail below with respectto FIGS. 17A and 17B. The ADC 106 can have 12 bits of SNR and −72 dBc oflinearity for full-scale signals.

The television receiver 100 does not include a SAW filter. Rather, thetelevision receiver 100 uses distributed filtering, in both the analogand digital domains, to isolate a desired television channel signal.This approach allows for the complete realization of the televisionreceiver 100 on a single Integrated Circuit (IC) as well as being ableto implement the television receiver 100 using reduced performanceconstraints for various processing components when a SAW filter is notused. Accordingly, filters can be implemented with reduced order andreduced Q, thereby requiring less area when realized on an IC.Furthermore, such a design is less sensitive to process variations andoperating conditions that vary, such as temperature and voltage.However, in some alternative embodiments a SAW filter can also be used.

Referring now to FIG. 3, shown therein is a block diagram of anexemplary embodiment of the RF processing block 102. The RF processingblock 102 includes an antenna 120, a low noise amplifier (LNA) 122, afirst variable gain amplifier (VGA) 124, a power meter 126, a mixingstage 128 having a mixer 130 and a frequency synthesizer 132, and asecond variable gain amplifier (VGA) 134. The power meter 126 isoptional depending on the automatic gain control method that is employedwhich is described in further detail below. The RF processing block 102receives and processes the wideband television signal 10 to provide amulti-channel television signal 136 that can include on the order oftens of television channel signals including the desired televisionchannel signal 22. Accordingly, the RF processing block 102 provides afirst level of filtering, as well as gain or attenuation, as the casemay be. In alternative embodiments, a cable connection, a satellite dishor other wireless connection can be used instead of the antenna 120.

The wideband television signal 10 is received by the antenna 120 andthen amplified by the LNA 122 and the VGA 124. The amount of gain thatis provided by the LNA 122 and the VGA 124 is variable based on certainproperties of the wideband television signal 10, and the desiredtelevision channel signal, which can vary widely. However, in some casesthe received television signal 10 may have to be attenuated. In general,the combined amount of gain provided by the LNA 122 and the VGA 124 canvary from −20 dB to 50 dB. Accordingly, both of the amplifiers 122 and124 have a wide dynamic amplitude range. The amount of gain provided bythe VGA 124 is controlled by AGC control signal, 138. Although signal138 is shown as analog, it can be either analog or digital.

The frequency response of the LNA 122 and the VGA 124 can extend up toabout 1 GHz to pass the received television signal along with someharmonics without significant distortion. However, in alternativeembodiments, at least one additional filter (i.e. a switchable filter, atracking filter, an FM notch filter, or an FM band filter for improvedFM performance) can be inserted before the LNA 122 to remove unwantedsignals such as cell phone and short-wave radio signals. Furthermore, FMradio signals can also be filtered out. Alternatively, FM radio signalscan be included if radio functionality is also desired. In alternativeembodiments, the LNA 122 and the VGA 124 can be combined and implementedin one variable gain stage.

The mixing stage 128 mixes the output of the VGA 124 to a much higherfrequency range on the order of GHz. At this higher frequency range, itis easier to implement the components of the analog processing block104. In addition, harmonics of the mixing stage 128 are at very highfrequencies thus avoiding any potential overlap with the television bandthat could result in any interfering images. Also, in this frequencyrange interference from other television signals can be minimized due tovarious signal processing techniques used in the analog signalprocessing block 104. For example, the mixing stage 128 can provide ahigh-side mix to mix the output of the VGA 124 so that the desiredtelevision channel signal 22 is centered near 1.125 GHz (see FIG. 5A).To achieve this, the frequency synthesizer 132 is tunable and, in somecases, can provide a tunable signal with a frequency in the range of1.16 GHz to 1.9 GHz. For example, with an oscillation frequency of 1.167GHz, the low end of the television band (i.e. −42 MHz) appears at 1.125GHz, and with an oscillation frequency of 1.987 GHz, the high end of thetelevision band (i.e. −862 MHz) appears at 1.125 GHz. The frequencysynthesizer 132 receives a tuning control signal 140 from the digitalprocessing block 108 based on the frequency of the desired televisionchannel signal 22 and the frequency at which the frequency content ofthe desired television channel signal is to be placed for processing bythe analog processing block 104.

The frequency synthesizer 132 acts as a Local Oscillator (LO) and can bebased on a PLL design and the oscillation signal can be derived from acrystal oscillator having a frequency of 4 MHz or 16 MHz for example.The frequency synthesizer 132 will have a certain amount of offset errorthat will need to be resolved by the digital processing block 108 tolocate the desired television channel signal. However, due to coarsefiltering that is provided in the analog processing block 104 and thesignal processing provided by the digital processing block 108, thefrequency synthesizer 132 can be realized with a coarser or larger stepsize such that there is a larger amount of offset in the exact locationof the desired television channel signal after mixing (i.e. a largershift from being centered at 1.125 GHz). Generally, the televisionreceiver 100 can tolerate offsets of 1 MHz or more, particularly whenthe offset is due to the use of a coarse synthesizer step size, sincethe offset in this case is known and can be compensated furtherdownstream in the processing chain. This is discussed in further detailbelow.

The mixing stage 128 can also provide some gain, and filtering. In someimplementations, the mixer 130 can include a differential gain stagethat can provide about 10-20 dB of gain, and can include a reactiveload, such as a pair of LC tank filters, to provide filtering around1.125 GHz. The filtering is not sharp since the inductors are realizedon an IC, and the Q of the LC tank filters can range from 6 to 12 atcertain frequencies. Accordingly, the bandwidth of the filteringprovided in the mixing stage 128 can be on the order of one hundred MHz,and the output of the mixing stage 128 can include over ten televisionchannels. Alternatively, bond-wires can be used for the inductors toachieve Q-values of up to 10 or greater. In at least some cases,external inductors can also be used for greater selectivity.

The output of the mixing stage 128 is then amplified by the VGA 134,which can be a standard VGA. The VGA 134 can generally be used toprovide about 10 to 30 dB of gain. The amount of gain provided by theVGA 134 is selected based on the gain provided by the VGA 124, as wellas the amount of gain and filtering that is provided by the mixing stage128. Gain control for the VGAs 124 and 134 is described in furtherdetail below. In alternative embodiments, if there is no filtering afterthe VGA 124 or if no additional gain is necessary, then the VGA 134 isoptional and can be excluded.

Referring now to FIG. 4, shown therein is a block diagram for anexemplary embodiment of the analog processing block 104. The analogprocessing block 120 generally filters and amplifies a multi-channeltelevision signal 136 to produce a coarse channel signal. The filteringis referred to as coarse filtering in that bandwidths are used for thefilters that are large enough to accommodate different bandwidths forthe desired television channel signal due to the various different typesof analog and digital broadcast standards that are used. The bandwidthsare also selected to be wide or large enough to accommodate anyfrequency shifts or offsets in the desired television channel signal aswell as any variability due to analog circuitry. This variabilityincludes component tolerances, temperature and voltage variations (whichcan result in a variation in absolute frequency and the bandwidth of thesignal path), irregularities which occur near the band edges of thefilters (using a wide band minimizes the effects when not pushing theedges) and easing the difficulty of precise analog design (morespecifically, the tuning required to set and keep the band pass of thefilters precise): These variations in frequency can also be tolerated bytracking the carrier frequency of the desired television channel signalfor both analog and digital broadcast standards. For analog broadcaststandards, the carrier tracking is akin to tracking the picture carrierand audio carrier for the desired television channel (although in someembodiments audio carrier tracking can be slaved to picture carriertracking as discussed below with regards to FIG. 13B). For digitalbroadcast standards, the carrier tracking is optional but when performedis akin to tracking a specified frequency such as a center frequency.For instance, even though DVB-T signals can be considered to have asmany as 8192 carriers, tuning can be done by specifying a centerfrequency for the 8192 carrier frequencies and tracking the middlecarrier frequency. The carrier frequency tracking is discussed furtherwith regards to FIGS. 10, 12A, 13A, 13B, 17A and 17B. When performed,the carrier tracking is used to introduce a frequency shift feedbacksignal to ensure that the desired television channel signal remains inthe bandwidth of the filtering components in the digital processingblock 108. This is described in more detail with regards to FIG. 10.

The analog processing block 102 includes a first coarse bandpass filter150, a third VGA 152, a sample and hold circuit 154, a discrete-timecoarse bandpass filter 156 (which can be based on a switched-capacitorimplementation), a discrete-time VGA 158 and a frequency synthesizer160. Gain control for the VGAs 152 and 158 is discussed in more detailfurther below. In some cases, one of the VGAs 152 and 158 is optionaland can be excluded in alternative embodiments. If the coarse bandpassfilter 150 is not present, then the VGA 152 is not required. If thediscrete-time bandpass filter 156 is not required, then the VGA 158 willnot be necessary. In addition, the coarse bandpass filter 150 can beimplemented in a discrete or integrated fashion. In an alternativeembodiment of the analog processing block 102, if a continuous-timebandpass sigma-delta ADC is used, then the sample-and-hold circuit 154is not required and the ADC has inherent anti-alias filteringpotentially precluding the need for other filters. Also the filter 156is a continuous-time filter and the VGA 158 is a continuous timevariable gain amplifier.

Generally, the analog processing block 104 processes the multi-channeltelevision signal 136 to provide a coarse channel signal 162 thatincludes the frequency content of the desired television channel signal22 as well as portions, or the entirety, of one or more adjacenttelevision channel signals. In some implementations, the coarse channelsignal 162 can have a bandwidth in the range of 10-20 MHz and in somecases can be approximately 10 MHz, and therefore, for some televisionbroadcast standards can include one full channel and two partialchannels (see FIG. 5B for example) or two full television channelsignals. The analog processing block 104 utilizes distributed, coarsefiltering to provide the coarse channel signal 162 with enough bandwidthto address the various issues mentioned previously. This is described infurther detail below.

The coarse bandpass filter 150 provides another level of filtering toremove unwanted television channel signals as well as to preventaliasing due to subsequent discrete time sampling. The coarse bandpassfilter 150 is also approximately centered at the frequency to which themixing stage 128 mixes the desired television channel signal. The sizeof the passband of the coarse bandpass filter 150 is large enough topass a coarse frequency region of interest 170 c (see FIG. 5C), whichincludes the desired television channel signal and at least portions ofat least one or more adjacent television channel signals.

The sampling rate of the sample and hold circuit 154 and the bandwidthof the coarse bandpass filter 150 can be selected to employ sub-samplingso that another mixer is not needed to shift the coarse frequency regionof interest to IF. The sampling rate is selected to be more than twicethe bandwidth of the coarse frequency region of interest 170 c, whichcan generally be about 10-20 MHz wide. However, it is not practical forthe coarse bandpass filter 150 to have a bandwidth of 10-20 MHz with acenter frequency in the Gigahertz range. Rather, the sampling rate ofthe sample and hold circuit 154 can be set much higher than twice thebandwidth of the coarse frequency region of interest 170 c, which allowsthe passband, transition band, and stopband requirements of the coarsebandpass filter 150 to be relaxed. Further, the amplitude response ofthe coarse bandpass filter 150 in the passband also does not have to beflat since it can be corrected digitally as is further described belowwith relation to FIG. 9. However, in other embodiments other types ofsampling can be used such as direct sampling or over-sampling ratherthan sub-sampling.

The amount of attenuation provided by the coarse bandpass filter 150 ischosen in relation to the sampling frequency used by the sample and holdcircuit 154, the bandwidth of the frequency region of interest 170 c,and the amount of resolution required for the ADC 106. For example,assume that the sample and hold circuit 154 uses a sampling rate of 500MHz, the passband of the coarse bandpass filter 150 is about 500 MHz,and the mixing stage 128 mixes the desired television channel signal to1.125 GHz. This sub-sampling, otherwise known as undersampling, placesan image of the desired television channel signal that was originally inthe 1.125 GHz region at 125 MHz, and other images 170 a and 170 b of theoutput signal 170 of the coarse bandpass filter 150 centered atmultiples of 250 MHz away from 125 MHz as shown in FIG. 5C. As can beseen, the attenuation provided by the coarse bandpass filter 150 is suchthat the sub-sampled version of its output 170 overlaps with images 170a and 170 b. However, the attenuation provided by the coarse bandpassfilter 150 is selected such that at the region of overlap between thesub-sampled version of the coarse frequency region of interest 170 c andthe skirts of the images 170 a and 170 b, the skirts of the images 170 aand 170 b are sufficiently attenuated to provide an adequate amount ofresolution without aliasing when the ADC 106 samples the frequencyregion of interest. For example, in at least some cases, the amount ofattenuation can be at least −74 dB at +/−BW/2 from 125 MHz to ensurethat the ADC 106 will have 12-bit resolution where BW/2 is half of thebandwidth of the coarse bandpass filter 150. It should be noted that asmaller bandwidth can be selected for the coarse bandpass filter 150 ifinductors with a higher Q are employed. For example, Q-enhancedinductors, bond-wires, and the like can be used to increase the Q to 5and reduce the bandwidth to about 250 MHz or so.

In some implementations, the coarse bandpass filter 150 can be realizedas a 6^(th) order filter (see FIG. 6A) which can be implemented using acascade of three 2^(nd) order LC tank filter stages, similar to the LCtank filters used in the mixing stage 128. Each tank filter stage can beseparated by a buffer to avoid interaction with one another.Furthermore, each buffer can be used as a distributed source of gaincontrol to help maintain a reasonable signal level as the signal powerdecreases with each subsequent LC tank filter stage. The overall Q ofthe bandpass filter 138 can be about 12. The coarse bandpass filter 150can be implemented in other ways, as is commonly known by those skilledin the art.

The sample and hold circuit 154 is provided with a timing signal by thefrequency synthesizer 160 to perform sub-sampling. The frequencysynthesizer 160 receives a timing control signal 166 from the digitalprocessing block 108 so that the rate of sub-sampling can be varied ifneeded. An aperture window is associated with the sample and holdcircuit 154, and the length of the aperture window can be selected suchthat the multi-channel television signal 136 can be resolved to theleast significant bit of the ADC 106. It should be understood that allblocks following the sample and hold circuit 154 are implemented withdiscrete-time components.

The coarse bandpass filter 156 can be used to provide another level offiltering. The coarse bandpass filter 156 can be realized as a switchedcapacitor filter with a passband centered at 125 MHz, for this example.The coarse bandpass filter 156 similarly provides a coarse channelsignal output that includes the desired television channel signal.However, the coarse bandpass filter 156 has a sharper transfer functionthan the coarse bandpass filter 150 in that it provides a larger amountof attenuation (i.e. the rolloff is larger) to deal with the moreextreme blocking profiles in which the adjacent television channelsignal may be 35 to 40 dB larger than the desired television channelsignal (this depends on the television broadcast standard). In someimplementations, the coarse bandpass filter 156 can be realized as an8^(th) order filter (see FIG. 6B) with a much steeper rolloff than thatof the coarse bandpass filter 150 to further improve the signal-to-noiseratio (SNR) for the desired television channel signal. In some cases,the coarse bandpass filter 156 can also be used to limit the power ofthe coarse channel signal 162 such that the resolution of the ADC 106 issufficient to digitize this signal and to resolve the desired televisionchannel signal to the necessary accuracy.

The VGAs 152 and 158 can provide an appropriate amount of gain to theoutput of the coarse bandpass filters 150 and 156, respectively, basedon the amount of filtering that was done by these filters and the levelof the received signal. Furthermore, due to the diverse RF signalblocking profiles (one of which is shown in FIG. 1B for example),various gain combinations can be selected to yield improved performance.Accordingly, the VGAs 124, 134, 152 and 158 receive gain control signals138, 142, 164 and 168 from the digital processing block 108 to moreeffectively apply a distributed amount of gain or attenuation.

The digital processing block 108 determines the optimal distribution andthe amount of gain/attenuation. The digital processing block 108 candetermine gain distribution in a number of ways. For instance, thedigital processing block 108 can determine gain distribution based onmeasurements that are made by the power meter 126. In this case, thepower meter 126 provides analog signal information 144 to the digitalprocessing block 108 (this can be provided in a digital manner).Alternatively, the digital processing block 108 can use other methodsfor controlling the gain of the RF and analog processing blocks 102 and104 in which case the power meter 126 can be optional.

Generally, the amount of gain provided by the RF processing block 102and the analog processing block 104 can be adjusted to affectsensitivity and distortion of signals at various locations in the RF andanalog processing blocks 102 and 104. The amount of gain provided by theanalog processing block 104 can be adjusted such that the input range ofthe ADC 106 is fully utilized while controlling the amount of distortionin blocks 102 and 104. In some cases, the entire amount of gain providedby the RF and analog processing blocks 102 and 104 can be on the orderof 100 dB. Gain control techniques that can be used are described inmore detail below with reference to FIGS. 15, and 16A-16C.

The ADC 106 digitizes the output of the analog processing block 104 toprovide a digitized coarse channel signal 172. The digitized coarsechannel signal 172 includes the desired television channel signal, andportions of one or more adjacent television channel signals depending onthe television broadcast standard. The coarse filtering provided by thevarious components in the RF processing block 102 and the analogprocessing block 104 dictate the number of bits that are required forthe ADC 106. If the mixing stage 130 and the coarse channel filters 150and 156 provide more filtering, then the ADC 106 can be implemented witha smaller number of bits. Accordingly, selecting the filteringcharacteristics of these components represents a balance between thecomplexity of providing greater selectivity in filtering versus the needto provide higher resolution in the ADC 106.

The ADC 106 can be realized with a bandpass delta sigma (BDS) ADC thatcan provide 11 or 12 effective bits (1 bit provides about 6 dB ofresolution/gain for the sampled signal). The BDS ADC oversamples at avery high rate with 3-4 actual bits to produce 11 or 12 effective bitsfor digitization. The BDS ADC outputs the data at an IF frequency.However, in other implementations it may be possible to realize the ADC106 with a 12-bit Nyquist rate ADC.

In alternative embodiments, the sample and hold circuit 154 and thecoarse bandpass filter 156 can be realized with a continuous-time filterand the VGA 158 with a traditional continuous-time VGA if acontinuous-time bandpass sigma-delta converter is used for the ADC 106.In some cases, a lowpass ADC can be used provided that the appropriatefiltering in the analog processing block 104 precedes the lowpass ADC.The input intermediate frequency can also be altered.

Also, in alternative embodiments, depending on the frequency range ofthe desired television channel signal as well as any mixing or otherfrequency shifting that is employed by the RF and analog processingblocks 102 and 104, it is possible to replace at least some of thecoarse bandpass filters described for the analog processing block 102with coarse lowpass filters.

Referring now to FIG. 7, shown therein is a block diagram of anexemplary embodiment of the digital processing block 108. The digitalprocessing block 108 includes an input filtering block 180, a videoprocessing block 182, first and second audio filtering blocks 184 and186, and an audio processing block 188. The digital processing block 108also includes a control block 190 for controlling various blocks in thedigital processing block 108 as well as providing timing and controlsignals to various components in the RF and analog processing blocks 102and 104. Certain portions of the digital processing block 108 operateaccording to the sampling rate used by the sample and hold clock circuit154; however, other sampling rates are also used by employingdownsampling or interpolation. The digital processing block 108 isgenerally configured to operate in an analog operation mode forprocessing signals transmitted according to an analog broadcaststandard, or a digital processing mode for processing signalstransmitted according to a digital broadcast standard.

The digital processing block 108 generally processes the digitizedcoarse channel signal 172 to recover the video and audio information forthe desired television channel signal. The processing takes into accountthe television broadcast standard used to transmit the desiredtelevision channel signal. The digital processing block 108 can producevarious outputs, depending on the particular implementation. Theseoutputs generally include, in different embodiments, variouscombinations of: digitized versions of a CVBS (Composite Video BasebandSignal) output for PAL/SECAM/NTSC formats, left and right channel audiooutputs, a sound IF output that can be further processed by an audiodecoder (not shown), and a digital video output, which can be furtherprocessed by a digital television demodulator to provide a digitaltransport stream that can then be processed by another element such asan MPEG-2 decoder, for example. The digital transport stream includescompressed digital data representing the audio and video information ofone or more television programs. Additionally, a digital IF output maybe provided to an external digital demodulator. The digitized versionsof the CVBS, baseband audio, sound IF and digital IF outputs may beconverted to analog form using the DAC block 110. In alternativeembodiments, these signals may be conveyed in digital form to subsequentprocessing stages without need for further conversion.

More particularly, the digitized coarse channel signal 172 is processedby the input filtering block 180 to provide a processed digitized coarsechannel signal 192. The input processing block 180 generally provides acombination of down conversion, pre-filtering and downsampling. Theinput filtering block 180 also includes components for setting the gainof various variable gain amplifiers in the RF and analog processingblocks 102 and 104. However, in alternative embodiments thisfunctionality can be provided by the control block 190. The processeddigitized coarse channel signal 192 is then processed by the videoprocessing block 182, which provides output signals 194 according to adesired output format. The output signals 194 include only videoinformation if the television broadcast standard is analog. However, ifthe television broadcast standard is digital, then the output signals194 include video and audio content in a format that is modulatedaccording to the broadcast standard. Further, the video processing block182 has two modes of operation: an analog operation mode to process thedesired television channel signal 22 when it is transmitted according toan analog broadcast standard, and a digital operation mode to processthe desired television channel signal 22 when it is transmittedaccording to a digital broadcast standard. This is described in furtherdetail below.

If the desired television channel signal 22 is transmitted according toan analog broadcast standard, then the processed digitized coarsechannel signal 192 is also processed by at least one of the first andsecond audio separation blocks 184 and 186 which provide at least one ofintermediate audio signals 196 and 198. In alternative embodiments, asexplained in further detail below with relation to FIG. 13B, instead ofreceiving signal 192, the first and second audio filtering blocks 184and 186 can receive a processed version of this signal provided by thevideo processing block 182. The first and second audio filtering blocks184 and 186 can also provide sound IF signals SIF1 and SIF2. Theintermediate audio signals 196 and 198, and the sound IF signals SIF1and SIF2 are then processed by the audio processing block 188 to providean audio output signal 200 according to a desired output format. Thefirst and second audio separation blocks 184 and 186 may both be usedfor situations in which the analog television broadcast standarddictates the use of two audio carriers. Alternatively, in this case, asingle audio separation block can be used to separate both carriers atthe same time and to provide them via a single SIF connection to adownstream audio device for further processing. For analog broadcaststandards that utilize a single audio carrier, only one of the first andsecond audio separation blocks 184 and 186 are enabled by the controlblock 190.

In at least some cases, the control block 190 can also receive signalsB1 and B2 from the video processing block 182 and a digital demodulatorrespectively to determine whether the television broadcast standard usedfor the desired television channel signal is analog or digital. Forinstance, the video processing block 182 can be first operated in theanalog operation mode and if a lock to the picture carrier is achievedin a reasonable amount of time then this lock is identified via thesignal B1 (see FIG. 12A) so that the control block 190 can configure thevarious components of the digital processing block 108 for analogoperation mode. If a lock is not obtained, then the control block 90 canconfigure the video processing block 182 for operation in the digitaloperation mode and determine if the desired television channel signal isproperly demodulated. In this regard, a component of a digitaldemodulator (see FIG. 17B) can indicate successful demodulation in thesignal B2.

The control block 190 also determines the particular type of analog ordigital television broadcast standard. One way to determine thetelevision broadcast standard is to detect the type of audio informationthat is included in the processed digitized coarse channel signal 192.Alternatively, the television receiver 100 can be configured for aparticular broadcast standard. In this respect, the architecture of thetelevision receiver 100 allows the receiver 100 to be mass-produced andthen configured, by setting certain parameters, for processingtelevision signals transmitted according to a certain broadcaststandard.

Referring now to FIG. 8, shown therein is a flow chart diagram of anexemplary embodiment of a standards detection method 250 which can beused for detecting the television transmission broadcast standard thatwas used to transmit the wideband television signal 10. The standardsdetection method 250 includes a digital detection mode for detectingwhether the desired television channel signal is a digital televisionsignal and an analog detection mode for detecting whether the desiredtelevision channel signal is an analog television signal. This includesreceiving feedback from the video and audio processing blocks 182-188.Once the bandwidth of the desired television channel signal is found,the control block 190 sets the bandwidth (BW) and the mode of operationfor the digital processing block 108.

At step 252, the standards detection method 250 enters digital detectionmode to determine whether the desired television channel signal wastransmitted with a digital broadcast standard. A first digital broadcaststandard is selected and the method 250 attempts to demodulate anddecode the desired television channel signal according to the selecteddigital broadcast standard using methods known by those skilled in theart. If a lock is obtained, then the method 250 moves to step 256 andsets the bandwidth used in various blocks in the video processing block182 during digital reception mode. If a lock is not obtained, the method250 moves to step 258 to determine whether there are any other digitalbroadcast standards to check. If so, another digital broadcast standardis checked and the method 250 moves back to step 254. If there are nomore digital broadcast standards to check, the method 250 moves to step260.

At step 260, the method 250 enters analog detection mode. At step 262,the method 250 attempts to lock to a carrier frequency. If a carrierfrequency is locked, the method 250 must confirm that the carrierfrequency is a picture carrier frequency and not an audio carrier.Several techniques can be used to lock to a carrier frequency andconfirm that it is a picture carrier. One technique includes using acoarse version of sync decoding to determine if sync information isassociated with the locked carrier frequency. If a lock has been made toa picture carrier frequency, the method 250 then moves to 264 otherwisethe method 250 continues to attempt to lock to a picture carrierfrequency. The coarse sync detection method involves counting the linesper field to verify that the locked carrier is a picture carrier.

At step 264, the method 250 attempts to locate the audio carrier. Toaccomplish this, a look-up table may be consulted which includes thepossible positions of the audio carrier based on the analog broadcasttransmission standard. The method 250 can start with the audio carrierthat is positioned furthest away from the picture carrier frequency todetermine if an audio carrier exists at that position. If a signal isfound at the audio carrier frequency then it is assumed to be an audiosignal but can be checked to make sure that it is not another picturecarrier by performing the coarse sync decoding. If an audio carrier isnot located, then the method looks at the audio carrier that is secondfurthest from the located picture carrier and repeats this processiteratively until detecting the audio carrier. If the audio carrier isdetected, the method 250 can search for a second audio carrier sincesome analog broadcast standards employ two audio carriers. Once thesingle or dual audio carriers are detected, the method 250 moves to step256 to set the BW and the mode of operation for the digital processingblock 108. The BW can be set to be the difference between the picturecarrier frequency and the audio carrier frequency minus a few hundredKHz, so as to ensure that the audio signal is adequately attenuated inthe video path.

If the audio carrier is not located at step 264, then the method 250moves to step 266 to search for the next picture carrier signal. Thesearch range for the next picture carrier signal can be the widestbandwidth and guard band that is used in analog broadcast transmissionstandards. If no picture carrier signal is located, then the BW can beset to the maximum bandwidth for analog broadcast transmissionstandards. If another picture carrier is found, then the bandwidth canbe determined based on the distance of the newly located picture carrierfrequency compared to the initially located picture carrier frequency.

Based on the detected television broadcast standard for the desiredtelevision channel signal, the control block 190 provides a videocontrol information signal 202 to the video processing block 182 tocontrol the operation mode of the video processing block 182. Thecontrol block 190 also provides audio control information signals 204,206 and 208 to the first and second audio separation blocks 184 and 186,and the audio processing block 188. Generally, the audio controlinformation 204, 206 and 208 enables blocks 184, 186 and 188 when thetelevision broadcast standard used for the desired television channelsignal is analog. In this case, the audio control information 204 and206 enable at least one of the audio blocks 184 and 186 depending on thenumber of audio carriers in the processed digitized coarse channelsignal 192. The audio control information 204, 206 and 208 can includeother operational parameter values that are discussed in more detailbelow. The control block 190 also provides control and timinginformation signals 210 to the RF and analog processing blocks 102 and104. This information is derived from input information 212 based on thetelevision channel that a user of the television receiver 100 wishes toview. The control and timing information 210 is used to control thefrequencies of various synthesizers as well as sampling rates.

Referring now to FIG. 9, shown therein is a block diagram of anexemplary embodiment of the input filtering block 180. The inputfiltering block 180 includes a frequency rotator (i.e. a down-converteror digital mixer) 300 which receives a rotation control signal 302, adecimation filtering block 304, an equalizer 306, and an analog gaincontrol block 308. It should be noted that in alternative embodiments,the analog gain control block 308 can be located in a different block.Further in some alternative embodiments, the functionality of the analoggain control block 308 can be provided by the control block 190.

The frequency rotator 300 processes the digitized coarse channel signal172 by performing down conversion to the baseband such that the coarsefrequency region of interest 170 c is now centered about DC. The amountof down conversion is controlled by the rotation control signal 302,which is provided by a free-running discrete time oscillator (notshown). The choice of a sampling rate that is 4 times the centrefrequency of the range of interest allows the rotator 300 to beimplemented without the need for multipliers, since rotation occurs in90 degree increments. The frequency rotator 300 generates quadraturesignals, i.e. in-phase (I) and quadrature (Q) signals for improvedprocessing efficiency. The I and Q signals are then processed byseparate I and Q signal paths as is commonly known by those skilled inthe art. It should be noted that only one signal path withdouble/thicker lines are shown to denote the I and Q signal paths tosimplify the description. However, it should be understood that blockswith two I and Q inputs and two I and Q outputs are actually implementedwith two blocks; one block processes the I signal and the other blockprocesses the Q signal. This nomenclature is used in other figures aswell.

The decimation filtering block 304 provides some low pass filtering anddownsampling to the output of the frequency rotator 300 to removeunwanted signal components and to reduce the sample rate in order tosimplify subsequent processing stages. The filtering removesquantization noise that results from the digitization provided by theADC 106. If a bandpass sigma delta converter is used for the ADC 106,then the filtering can be designed to attenuate the noise-shapedspectral regions of the I and Q output signals of the frequency rotator300. The decimation filtering block 304 then performs downsampling sothat the other blocks in the digital processing block 108 can operatemore efficiently. For example, downsampling can be done so that thesampling rate associated with the processed digitized coarse channelsignal 192 is at approximately 31.25 MHz. The amount of low passfiltering provided by the decimation filtering block 304 is alsocommensurate with the amount of downsampling as is commonly known bythose skilled in the art.

The output of the decimation filtering block 304 is then processed bythe equalizer 306. It should be noted that the equalizer 306 is optionaland is not needed if the equalization functionality can be provided byanother downstream element; for example, equalization can be provided bya digital demodulator (not shown) that receives the output signal 194 ofthe video processing block 182. The functionality of the equalizer 306may also be optional when processing television signals that aretransmitted according to an analog broadcast standard, depending on thelevel of performance required for the output signal 194. Accordingly, insome cases the equalizer 306 can be disabled or not included.

The equalizer 306 processes the I and Q output of the downsampler tocompensate for the non-ideal filtering provided by the filters 152 and156 in the analog processing block 104. Accordingly, the equalizer 306provides equalization to make it seem as if the filters 152 and 156 havea flat inband response and improved group delay response. The equalizer306 can combine the I and Q signals to provide a single-ended realoutput signal 192. Alternatively, in some alternative embodiments, theequalizer 306 can provide I and Q output signals that are thenappropriately processed by the other blocks in the digital processingblock 108.

The equalizer 306 can be implemented in a variety of fashions as iscommonly known by those skilled in the art. For instance, the equalizer306 can include a real or imaginary FIR filter to perform equalization,and can also include several cascaded filters as well as a frequencyrotator to shift the signal by a certain fraction of the sampling rateso that the filters in the equalizer 306 can be implemented moreefficiently. In this case, the blocks 182, 184 and 186 include acorresponding frequency rotator to shift the processed digitized coarsechannel signal 192 back to the baseband.

The output of the ADC 106 is also provided to the analog gain controlblock 308. The purpose of the analog gain control block 308 is to adjustthe gain settings at the RF and IF stages to adjust the level of thesignal presented to the ADC 106 to ultimately increase the quality ofthe desired television channel signal. This may be performed using afeedback loop in which the level of the output from the ADC 106 ismeasured and compared to a preset reference level. If the measured levelis less than the reference level then the gain is increased, while ifthe measured level is higher than the reference level then the gain isdecreased. Therefore, the reference level may be thought of as a targetlevel which the loop seeks to maintain, even when the level of thesignal from the antenna 120 changes. The analog gain control block 308can perform this gain control technique, which is described in moredetail with regards to FIGS. 16A-16C.

The analog gain control block 308 is implemented digitally and providesRF gain control signals 138 and 142 to the VGAs 124 and 134,respectively, in the RF processing block 102. The analog gain controlblock 228 also provides IF gain control signals 164 and 168 to the VGAs152 and 158, respectively, in the analog processing block 104 if theseamplifiers exist (note they are optional). The analog gain control block308 can perform the gain control method outlined in FIG. 15 or othermethods described in further detail below. In alternative embodiments, again control system may be implemented which utilizes informationprovided by the power meter 126, in which case the analog gain controlblock 308 also receives the analog signal information 144.

Referring now to FIG. 10, shown therein is a block diagram of anexemplary embodiment of the video processing block 182. The videoprocessing block 182 processes the processed digitized coarse channelsignal 192 by generally performing carrier frequency recovery,resampling and filtering. The video processing block 182 can alsoperform phase noise reduction by compensating for phase perturbationswhich includes both noise and systematic variation caused by spurs andthe like. As mentioned, the video processing block 182 operates in adigital operation mode or an analog operation mode depending on whetherthe desired television channel signal is transmitted according to adigital or analog television broadcast standard.

The video processing block 182 includes a first frequency rotator 350, avideo pre-polyphase filter 352 p, a first video polyphase filter 352, avideo resampling control block 354, a video filter 356, a digital VGA358, a multiplexer 360, a digital gain control block 362, a secondfrequency rotator 364, a picture carrier recovery block 366, an outputequalizer 388, a second video polyphase filter 368, an up-sampling block370 and a Digital to Analog Converter (DAC) 372. The video pre-polyphasefilter 352 p, first video polyphase filter 352, video resampling controlblock 354, and the video filter 356 can be considered to be a videofilter stage. The picture carrier recovery block 366 includes a carrierrecovery filter 374, a first phase rotator 376, a carrier recovery block378, an AGC filter 380, a second phase rotator 382, a Vestigial SideBand (VSB) filter 384, a third phase rotator 386, an overmodulationfilter 406, and an overmodulation magnitude detector 408.

The control block 190 provides a mode control signal 388 to controlwhether the video processing block 182 operates in the analog or digitaloperation mode. The mode control signal 388 is provided to themultiplexer 360 as a selection input to select which gain control signalis applied to the digital VGA 358. This is described in further detailbelow. The mode control signal 388 is also provided to the picturecarrier recovery block 366 to enable this block during the analogoperation mode or to disable this block during the digital operationmode.

In both digital and analog operation modes, the frequency rotator 350generally shifts the frequency content of the processed digitized coarsechannel signal 192 so that the desired television channel is centeredabout DC. In the case of analog reception, the video informationincluding the vestigial sideband of the desired channel is centeredabout DC. In the case of digital reception, the entire channel issimilarly centered. Both of these cases are generally shown in FIG. 11A.The frequency rotator 350 also produces in-phase I and quadrature Qsignals, and the majority of the video processing block 182 includes twosignal paths for processing the I and Q signals. However, there is adegree of offset in the location of the frequency content of the desiredtelevision channel due to both known and unknown elements. The degree ofoffset can be larger for the analog operation mode compared to thedigital operation mode for some situations. A portion of the offsetresults from coarse positioning of the frequency content of the desiredtelevision channel signal due to using a coarse step size or a fine stepsize in the frequency synthesizer 132 and this amount of offset isknown. However, an additional offset arises from frequency tolerances inthe frequency synthesizer 132, and the transmitter, or other hardware,that transmitted the desired television channel signal 22 to thereceiver 100. These offsets are unknown and must also be corrected.

In the digital operation mode, the offset correction can be performed ata later stage by another component such as a digital demodulator (notshown). Those skilled in the art are familiar with techniques that canbe employed in the digital demodulator for correcting the offset fortelevision channel signals that are broadcast according to a digitalbroadcast standard. Accordingly, in some embodiments, there is nofeedback signal that is provided to the frequency rotator 350 duringdigital operation mode. Rather, the video processing block 182 canoperate in an open-loop or free-running fashion where a fixed or knownfrequency shift is applied to the processed digitized coarse channelsignal 192 to shift the signal to the baseband to attempt to center thecoarse frequency region of interest about DC in spite of the offset.Accordingly, known frequency offset errors in the carrier frequency,such as errors due to the use of a coarse step size or fine step size inthe oscillator used in the RF processing block 102 or due to any otherknown frequency offset errors, may be compensated in this way byapplying a corresponding known frequency shift via the frequency rotator350. The correction of other offsets that may be unknown can beperformed by a downstream digital demodulator (not shown).Alternatively, in other embodiments that include a digital demodulator(see FIGS. 17A and 17B for example), the digital demodulator can providea feedback signal to the frequency rotator 350 to adjust the shift thatis applied to the processed digitized coarse channel signal 192 so thatit is centered about DC regardless of the offset.

In the analog operation mode, the portion of the analog broadcasttelevision signal from frequency f₁ to f₂ should be centered about DC bythe frequency rotator 350. For example, FIG. 11B shows the location offrequencies f₁ and f₂ for an NTSC analog television channel signal as isused in North America. This is achieved by using a feedback loop, whichincludes the first and second frequency rotators 350 and 364 and thepicture carrier recovery block 366. The frequency rotator 350 applies avariable frequency shift to the processed digitized coarse channelsignal 192 to attempt to center the frequency content of the videoinformation of the desired television channel signal about DC. Uponinitial operation, the frequency rotator 350 applies an initialfrequency shift. The selection of the initial frequency shift isdiscussed in more detail with reference to FIG. 12A. The frequencyrotator 364 applies a fixed frequency shift of Δω such that the picturecarrier is shifted to DC when the frequency range f₁ to f₂ is centeredabout DC at the output of frequency rotator 350. Note that initially thepicture carrier may not be shifted exactly to DC due to the offseterror. The picture carrier recovery block 366 then tracks the frequencyoffset error by detecting the actual location of the picture carriersignal and generates an analog mode frequency shift feedback signal 390,which is provided to the frequency rotator 350 to adjust the amount offrequency shift that it provides. Over time, this adjustment infrequency shift is such that the fixed frequency shift provided by thefrequency rotator 364 moves the picture carrier signal to DC. Theoperation of the carrier recovery block 366 is described in furtherdetail below.

The output of the frequency rotator 350 is processed by the videopre-polyphase filter 352 p which provides low pass filtering to removeunwanted spectral components. The output of the video pre-polyphasefilter 352 p is processed by the video polyphase filter 352 whichprovides interpolation to change the number of data samples for theoutput of the video pre-polyphase filter 352 p. The interpolationoperation performed by the video polyphase filter 352 provides adifferent sampled version of the output of the video pre-polyphasefilter 352 p in that the data samples of the output of the videopolyphase filter 352 are at different temporal locations and have adifferent temporal spacing compared to the output of the videopre-polyphase filter 352 p. The end effect of the video polyphase filter352 is to change the sampling rate at the output of the video polyphasefilter 352 such that the bandwidth of the desired television channelsignal normalized with respect to the new sampling rate is transformedto match the bandwidth of the video filter 356. Accordingly, the videopolyphase filters 352 and 368 act as resampling elements. The amount ofinterpolation (i.e. the amount of resampling) is dictated by the videoresampling control block 354, which provides resampling control signals392 and 394 to the video polyphase filters 352 and 368 respectively. Thedegree of interpolation that is required is related to the broadcasttransmission standard that was determined by the control block 190,which is indicated by a broadcast information signal 286 that is derivedfrom the video control information 202 that is provided by the controlblock 190. In alternative embodiments, the functionality of the videoresampling control block 354 can be provided by the control block 190.

Consider the parameter W_(null)=f_(null)/f_(S) ^((polyphase output))which is the fixed cut-off frequency of the video filter 356 normalizedwith respect to the new sampling rate f_(S) ^((polyphase output)). Thevideo resampling control block 354 configures the value for the newsampling rate f_(S) ^((polyphase output)) such that f_(null) correspondsto half of the bandwidth of the desired television channel signal, sinceit is a complex signal centered about DC. The value for the parameterf_(null) varies based on the television broadcast standard. For analogtelevision broadcast standards, the desired television channel signalincludes the video information f_(v) above the picture carrier frequencyas defined by the particular broadcast standard, as well as thevestigial sideband portion f_(VSB) located below the picture carrierfrequency both in MHz and shown in FIG. 11B. Accordingly, f_(null) canbe given by the following formula.

f_(null)˜(f_(v)+f_(VSB))/2 MHz

The value of f_(VSB) is assumed to be 0.75 MHz in the followingexamples, though other values may also be possible. The value ofW_(null) can be set to 0.31 by design, for example, as a filter withthis characteristic represents a practical trade-off in designconsiderations. Some examples are shown below for various analogtelevision broadcast standards.

f_(null)˜(4.2+0.75)/2 MHz (NTSC)

f_(null)˜(5.0+0.75)/2 MHz (PAL B,G)

f_(null)˜(5.5+0.75)/2 MHz (PAL I)

f_(null)˜(5.7+0.75)/2 MHz (PAL D,K)

For digital television broadcast standards including DVB-T, thebandwidth of the desired television channel signal is the entire channelwidth as illustrated by the following examples (f_(VSB) is notapplicable). As in the previous examples, f_(null) is taken as half ofthis bandwidth since the signal is complex in nature.

f_(null)˜(6.0)/2 MHz (6 MHz DVB-T)

f_(null)˜(7.0)/2 MHz (7 MHz DVB-T)

f_(null)˜(8.0)/2 MHz (8 MHz DVB-T)

f_(null)˜(6.0)/2 MHz (6 MHz ATSC)

Accordingly, rather than implementing the video filter 356 as a variablebandpass filter, the video processing block 182 employs a fixedbandwidth for the video filter 356 and changes the effective samplingrate of the data provided to the video filter 356. The output of thevideo polyphase filter 352 is at the same physical clock rate but at anew effective sampling rate to adjust the spectrum or bandwidth of thedesired television channel signal to match the passband of the videofilter 356. This processing allows a sharp, fixed filtering block to beused as if it has a variable passband size. This results in a moreefficient implementation since coefficients for a variety of differentfilter transfer functions to match each of the television broadcaststandards is not needed. Rather just one set of coefficients for thevideo filter 356 are stored. This “resampling processing” allows thevideo processing block 182 to filter television channel signals havingdifferent bandwidths, such as 6, 7 or 8 MHz, with the same fixed filter.Further, it should be noted that the filtering provided by the videofilter 356, the equalizer 306, and the analog filters 150 and 156approximate the filtering provided by the SAW filter that isconventionally used in traditional television receivers. Decimationfiltering can also be used prior to the video filter 356 to reduce thenumber of data samples, and accordingly the number of coefficients usedfor the video filter 356.

The same filter coefficients may be used by the video filter 356 in thedigital and analog operation modes. In the analog reception mode, thesample rate is adjusted such that the video filter 356 separates thevideo information of the desired television channel signal including thevestigial sideband portion, as described earlier. In the digitalreception mode, the sample rate is adjusted such that the video filter356 passes the entire digital channel, since the video and audioinformation are transmitted together as a multiplexed data stream. FIG.11C shows an exemplary diagram for a desired television channelconforming to the ATSC digital broadcast standard as is used in NorthAmerica. In this case, the effective bandwidth may be set slightly widerthan the bandwidth of the desired television channel, in order that someoffset in the actual frequency of the received desired televisionchannel may be tolerated without the video filter 356 impinging on theband edges of the desired television channel. Although a slightly widerfilter bandwidth may help in some cases, it may harm performance if astrong adjacent channel is present. The narrow filter option may be usedwithout undesired effects in at least some embodiments by applyingfrequency offset feedback from the digital demodulator to the frequencyrotator 350 in order to keep the effective bandwidth of the video filter356 aligned with the desired television channel. The video filter 356can be implemented with a pipeline of smaller filters to make thefiltering process more efficient, as is commonly known by those skilledin the art.

The operation of the video pre-polyphase filter 352 p, video polyphasefilter 352 and the video filter 356 will now be described in greaterdetail with reference to FIGS. 11D-11L (the frequency domain isrepresented in actual frequencies (Hz) not in normalized frequencies inthese figures). The output of the frequency rotator 350 is firstprocessed by the video pre-polyphase filter 352 p. As shown in FIG. 11D,the output of the frequency rotator 350 includes the desired televisionchannel signal and out of band components. The video pre-polyphasefilter 352 p removes substantially all of the out-of-band andout-of-interest spectral components that would otherwise alias to thedesired television channel signal band after resampling by the videopolyphase filter 352. Accordingly, the video pre-polyphase filter 352 pattenuates the high frequency components of the output of the frequencyrotator 350. The frequency response of the pre-polyphase filter 352 p isshown in FIG. 11D by the dashed line. The response is flat within theband of the desired television channel signal. The stopband of thepre-polyphase filter 352 p specifies the subsampling ratio of thepolyphase filter 352. When the spectral components at the input of thepre-polyphase filter 352 p are greater than half of the sampling rate atthe output of the polyphase filter 352, those components are aliasedinto the output of the polyphase filter 352. Accordingly, the minimumpossible sampling rate at the output of the polyphase filter is limitedby the need to avoid this aliasing. These possible aliased componentsare shown by the dotted lines for the boundary case (i.e. the minimumpossible sampling rate after polyphase sampling f_(S)^((polyphase output min))).

The video polyphase filter 352 reduces the rate f_(S)^((polyphase input)) of the input data samples to the output samplingrate f_(S) ^((polyphase output)). The resampling ratio is f_(S)^((polyphase output))/f_(S) ^((polyphase input)). The video polyphasefilter 352 is able to change the sampling rate within a particular rangeas limited by a stopband boundary frequency F_(boundary)^((pre-polyphase filter)) of the video pre-polyphase filter 352 p and anupper boundary frequency of the desired television channel signalF_(boundary) ^((desired television channel signal)) as follows.

f _(S) ^((polyphase output)) −F _(boundary) ^((pre-polyphase filter)) >F_(boundary) ^((desired television channel signal))

The spectrum after the polyphase filter 352 in actual frequencies (Hz)is shown in FIG. 11E. Now the bandwidth of the desired televisionchannel signal matches the bandwidth of the main video filter 356 andthe desired television channel signal is not damaged by any aliasedsignals. Other television channel signals and out of band aliasedspectral components are at least partially attenuated by thepre-polyphase filter 352 p.

The output of the video polyphase filter 352 is processed by the videofilter 356. The purpose of this arrangement is that by changing thesampling rate, the effective bandwidth of the video filter 356 may bechanged even though the filter itself remains fixed. Video filter 356 isa non-variable low pass filter with a nearly rectangular frequencyresponse that has a very sharp transition between the passband and thestopband. The frequency response and bandwidth are constant in thenormalized frequency domain f/f_(S), but in the absolute frequencydomain they are effectively varied by changing the sampling rate. Theoutput sampling rate f_(S) ^((polyphase output)) is chosen such that theeffective passband of the video filter 356 matches the bandwidth of thedesired television channel signal and such that any undesired spectralcomponents are removed. For different television broadcast standards thedesired television channel signal may have a different bandwidth. Theoperation of the frequency rotator 350 centers the desired televisionchannel signal about DC and by controlling the output sampling ratef_(S) ^((polyphase output)), the effective bandwidth of video filter 356is modified to match the bandwidth of the desired television channelsignal.

The video filter 356 filters the output of the video polyphase filter352 to pass the frequency content of the desired television channelsignal while rejecting the portions of the adjacent television signalsthat were in the coarse frequency region of interest. The video filter356 extracts the desired television channel signal with high precision.The video filter 356 is a steep rectangular filter that is constant in anormalized frequency domain with respect to sampling rate. The frequencyresponse of the video filter 256 is shown in FIGS. 11E and 11F and theoutput of the video filter 256 is shown in FIG. 11F.

The digital VGA 358 then amplifies the output of the video filter 356.In the analog operation mode, the amount of digital gain provided by thedigital VGA 358 is dictated by the digital gain control block 362, whichprovides a digital gain control signal 398 to the digital VGA 358. Inthe digital operation mode, the amount of gain provided by the digitalVGA 358 is dictated by a downstream digital demodulator block (see FIGS.19A and 19B for an example), which provides a digital gain controlsignal 400 to the digital VGA 358. The mode control signal 388 isprovided to the multiplexer 360 to select the digital gain controlsignal 398 if the operation mode is the analog operation mode, and thedigital gain control signal 400 if the operation mode is the digitaloperation mode. The generation of these signals is described in furtherdetail below.

In the digital operation mode, the I and Q output signals from thedigital VGA 358 are provided as the output signal 194 of the digitalprocessing block 182. These I and Q signals can then be furtherprocessed as desired, for instance, by a downstream digital demodulator,or by other processing elements as is well known by those skilled in theart. The digital demodulator would provide a digital transport streamthat can then be operated on by another element such as an MPEG-2decoder to produce video.

In the analog operation mode, the I and Q outputs of the digital VGA 358are provided to the frequency rotator 364, which provides a frequencyshift so that the picture carrier signal for the video component of thedesired television channel signal is shifted to DC. To accomplish this,the frequency rotator 364 provides a fixed frequency shift Δω that isequal to the difference between the center frequency of the video filter356 and the expected location of the picture carrier frequency.

The I and Q output signals of the frequency rotator 364 are provided tothe picture carrier recovery block 366 to determine a value for theanalog mode frequency shift feedback signal 390 so that the frequencyrotator 350 can center the frequency content of the desired televisionchannel signal about DC. The control block 190 enables the picturecarrier recovery block 366 to operate in the analog operation mode viathe mode control signal 388. The picture carrier recovery block 366 usesthe picture carrier frequency to demodulate the analog video informationin the output of the frequency rotator 364 to provide a CVBS output asthe output signal 194.

More particularly, the output of the frequency rotator 364 is providedto the carrier recovery filter 374, the AGC filter 380, the VSB filter384, and the overmodulation filter 406. In alternative embodiments,these blocks can be implemented in one block for improved efficiency byreusing elements required for filtering such as registers. Furthermore,in some cases, some of the filters 374, 380 and 406 can be implementedusing the same filter coefficients as described below.

The carrier recovery filter 374 is a narrowband filter centered at DCwith a bandwidth that is wide enough to pass only the picture carriersignal in order to separate it from the rest of the desired televisionchannel signal (see FIG. 11B). The carrier recovery filter 374 filtersits input to produce a filtered picture carrier signal. The phase of thefiltered picture carrier signal is then rotated by the phase rotator 376to compensate for phase noise in the output of the frequency rotator364. The phase rotator 376 applies a phase adjustment to produce aphase-adjusted filtered picture carrier signal that is then provided tothe carrier recovery block 378. Phase noise and the operation of thecarrier recovery block 378 is described in more detail with relation toFIG. 12A. In alternative embodiments, the control block 190 can providea bandwidth control signal (not shown) to control the bandwidth of thisnarrowband filter. The bandwidth can be controlled during differentstages of the picture carrier recovery process such as the acquisitionstage versus the lock stage; this is discussed in more detail withreference to FIG. 12A.

The AGC filter 380 is employed to precondition the output of thefrequency rotator 364 which is a frequency-shifted version of thedesired television channel signal. The AGC filter 380 performspreconditioning to remove noise and spurious signals. In some cases, theAGC filter 380 can have the same filter coefficients as the carrierrecovery filter 374. The AGC filter 380 filters its input to produceanother filtered picture carrier signal, the phase of which is thenrotated by the phase rotator 382 to compensate for phase noise in theoutput of the frequency rotator 364. Accordingly, the phase rotator 382applies a phase adjustment to produce another phase-adjusted filteredpicture carrier signal, which is provided to the digital gain controlblock 362. Thus, during the sync pulses, when the desired televisionchannel signal is at a peak level, the digital gain control block 362can react to the peak level of the desired television channel signalrather than noise or signal spikes so that the digital gain controlblock 362 can properly adjust the amount of gain applied to the digitalVGA 358 via the digital gain control signal 398. Accordingly, thedigital gain control block 362 can include a sync detector to detect thesync pulses and adjust the value of the digital gain control signal 398.In other embodiments, the phase rotator 382 can be eliminated when thedigital gain control block 362 computes the magnitude of the output ofthe AGC filter 380 since the magnitude function is phase insensitive.

In addition, the gain provided by the digital gain control block 362 canbe adjusted on a line-by-line basis by detecting the magnitude of eachsync pulse. Furthermore, to determine the amount of gain that should beapplied by the digital VGA 358, a desired target level can be set. Thedifference between the desired target level and the detected peak levelof the desired television channel signal can be used to determine thevalue of the digital gain control signal 298 so that the differencetends to zero.

The overmodulation filter 406 is employed to remove noise and spurioussignals at the output of the frequency rotator 364. The magnitude leveldetector 408 then detects the magnitude of the I and Q components of theoutput of the overmodulation filter 406 and provides a magnitude levelsignal 402 to the carrier recovery block 378, the purpose of which isdescribed further below in relation to FIG. 12A.

The video signal from the output of the frequency rotator 364 isprocessed by the VSB filter 384. The VSB filter 384 filters the outputof the frequency rotator 364 (i.e. the frequency-shifted version of thedesired television channel signal) to produce filtered videoinformation, i.e. the video information of the desired televisionchannel signal. The output of the VSB filter 384 is processed by thephase rotator 386 and the applied phase correction value 404 iscalculated by carrier recovery block 378. The VSB filter 384 and thephase rotator 386 together convert the input complex vestigial sidebandsignal to a real CVBS video signal. Under the condition that carrierrecovery lock has been acquired, i.e. at the output of the phase rotator386, the picture carrier is shifted to DC and its phase is 0 so that itis completely a real signal, the CVBS signal can be obtained byinverting the sign of the imaginary part spectral components that havenegative frequencies, leaving the positive frequency componentsunchanged and adding both to the real part of the complex input signal.The spectrum of the complex input signal is shown in FIGS. 11G to 11Iwhile the frequency response of the VSB filter 384 is shown in FIGS. 11Jto 11L. Since the final phase rotation by the phase rotator 386 is doneafter the complex VSB filter 384, the VSB filter 384 has to generatereal and imaginary parts that are provided as an input to the phaserotator 386. The CVBS signal is the real output of the phase rotator386. To generate the real output, the VSB filter 384 multiplies theimaginary part of the input signal spectrum S_(VS) ^(Im)(f) by theimaginary part of the frequency response H_(VSB) ^(Im)(f) of the VSBfilter 384, the product of which is added to the product of the realpart of the input signal S_(VS) ^(Re)(t) and the real part of thefrequency response H_(VSB) ^(Re)(f) of the VSB filter 384. It should benoted that the real part of the frequency response H_(VSB) ^(Re)(f) ofthe VSB filter 384 is unity at all frequencies. To generate theimaginary output, the VSB filter 384 multiplies the real part of theinput signal spectrum S_(VS) ^(Re) (f) by the imaginary part of thefrequency response H_(VSB) ^(Im)(f) of the VSB filter 384, the productof which is added to the product of the imaginary part of the inputsignal S_(VS) ^(Im)(f) and the real part of the frequency responseH_(VSB) ^(Re)(f) of the VSB filter 384. The phase rotator 386 thenapplies a phase adjustment to the phase of the output of the VSB filter384 to produce phase-adjusted video information which has beencompensated for the phase noise in the output of the frequency rotator364. The “real-only” output of the phase rotator 386 is then provided tothe output equalizer 388, which applies group delay correction accordingto television broadcast standards for the desired television channelsignal. The output of the output equalizer 388 is then provided to thevideo polyphase filter 368 so that the signal can be re-interpolated tocorrespond with the original sampling rate. The output of the videopolyphase filter 368 is then up-sampled by the up-sampling block 370 andconverted into an analog form by the DAC 372 and provided as the outputsignal 194 which in this case is an analog CVBS TV output signal.

The carrier recovery block 378 provides the analog mode frequency shiftfeedback signal 390 to adjust the amount of frequency rotation providedby the frequency rotator 350. The adjustment is such that the frequencyshift provided by the frequency rotator 364 aligns the picture carrierat DC. The carrier recovery block 378 is similar to a phase-locked loopand is discussed further in relation to FIG. 12A. However, in thisexemplary embodiment, the carrier recovery block 378 also provides aphase control signal 404 to the phase rotators 376, 382 and 386 tocontrol the amount of phase rotation that is applied. A phase rotator isa complex mixer that adjusts the phase of its input to correct for phaseerrors. However, in some cases, these phase rotators can be optional.Accordingly, in alternative embodiments, the picture carrier recoveryblock 366 does not include phase rotators 376, 382 and 386 and thecarrier recovery block 378 does not provide the phase control signal404.

In this exemplary embodiment, the picture carrier recovery block 366performs frequency correction to account for frequency offset errors inthe processed digitized coarse channel signal 192 and phase noisecompensation to compensate for phase perturbations to reduce phase noisein the processed digitized coarse channel signal 192. The phase noisereduction includes compensating for phase perturbations which includesphase noise and systematic variations caused by spurs and the like.Typically, a television tuner imparts phase noise onto the incomingtelevision signal 10. The phase noise typically appears as white noisein the processed television signal. If the phase noise is less than 200kHz or so, the phase noise can be tracked and attenuated. Conventionaltelevision receivers do not track or compensate for phase noise. Itshould be understood that the phase noise reduction provided by thepicture carrier recovery block 366 can be used with other televisionreceivers that employ a different technique for filtering an inputsignal that must then be demodulated to obtain the desired televisionchannel signal. In these cases, the picture carrier recovery block 366can be applied to the filtered input signal and the picture carrierrecovery block 366 does not have to generate the analog mode frequencyshift feedback signal 390.

Accordingly, when operating in analog operation mode, there are actuallytwo processing loops in the video processing block 182. There is anouter loop including the frequency rotator 350, video pre-polyphasefilter 352 p, video polyphase filter 352, video filter 356, digital VGA358, frequency rotator 364, and the picture carrier recovery block 366that acts like a frequency tracking loop. There is also an inner loopincluding the phase rotators 376, 382 and 386, and the carrier recoveryblock 378 that acts as a phase tracking loop. The frequency trackingloop tracks the carrier frequency of the desired television channelsignal and corrects for frequency offset errors to lock onto the picturecarrier frequency of the desired television channel signal. The phasetracking loop tracks and reduces the phase noise in the desiredtelevision channel signal. The phase tracking loop requires a highbandwidth in order to react quickly to the phase noise. Accordingly,filters employed in the phase tracking loop have a reduced number offilter taps and other delays in this loop are kept at a minimum. Thefrequency lock loop cannot track the phase noise since it has too muchdelay due to the amount, and sharpness, of the filtering that is done.

The first frequency rotator 350, video pre-polyphase filter 352 p, firstvideo polyphase filter 352, video resampling control block 354, videofilter 356, digital VGA 358, and the second frequency rotator 364 can beconsidered to be elements of a signal processing pathway thatcumulatively provide frequency shifting and filtering to removeextraneous signal components and output down-shifted frequencycomponents of the desired television channel signal component includingthe picture carrier signal frequency. Also the video pre-polyphasefilter 352 p and the first video polyphase filter 352 can be consideredto be a video polyphase filter stage for resampling a signal at a newsampling rate. The picture carrier recovery block 266 then generallyreceives the down-shifted frequency components of the television channelsignal component, locks onto the picture carrier signal frequency, andprovides a demodulated television channel signal. In at least someembodiments, during the locking process, the operation of the picturecarrier recovery block 366 can be modified when operating in anovermodulation handling mode to deal with the presence of overmodulationwhen locking onto the picture carrier.

The carrier recovery block 366 modifies its operation in the presence ofovermodulation in the filtered picture carrier signal when tracking atleast one of a frequency error signal and a phase error signal byapplying a weight to at least one of the frequency error signal and thephase error signal or by using a previous correction value. The phaseerror signal is produced by comparing a phase of the filtered picturecarrier signal with a phase reference signal. There can be embodimentsin which only frequency tracking is employed by tracking the frequencyerror signal, only phase noise reduction is employed by tracking thephase error signal as well as embodiments in which both frequency andphase tracking is employed as shown in FIG. 12A. Accordingly, there canbe embodiments in which the operation of the picture carrier recoveryblock 366 is modified in the presence of overmodulation while generatingthe feedback signal 390 that is provided to the signal processingpathway. There can also be embodiments in which the operation of thepicture carrier recovery block 366 is modified in the presence ofovermodulation during the production of the phase control signal 404 toreduce phase noise. There can also be embodiments in which the operationof the picture carrier recovery block 366 can be modified duringovermodulation while producing both the feedback signal 390 and thephase control signal 404. The description that follows is for theembodiment in which overmodulation handling is used for both thegeneration of the phase control signal 404 and the feedback signal 390,but it can be modified by applying overmodulation handling to theproduction of only one of these two signals as mentioned.

Referring now to FIG. 12A, shown therein is a block diagram of anexemplary embodiment of the carrier recovery block 378 which generallyincludes a phase correction stage, a frequency correction stage and astatus stage. The phase correction stage processes the phase rotatedoutput of the carrier recovery filter 374 and the magnitude level signal402 to produce the phase control signal 404. The frequency correctionstage also processes the phase rotated output of the carrier recoveryfilter 374 and the magnitude level signal 402 to produce the analog modefrequency shift feedback signal 390. The status stage receives phase andfrequency error signals from the phase and frequency correction stages,respectively, to determine if there is phase lock and/or frequency lock.Each of these stages are now discussed in more detail.

The phase correction stage includes a cordic block 450, aphase-frequency detector 452, a lowpass filter 454, a phase adjustmentblock 456, a phase loop amplifier 458, and a phase oscillator block 460.The phase adjustment block 456 includes a phase inversion detector 462,and a phase inversion block 464. The phase oscillator block 460 includesa phase accumulator 466 and a cordic block 468.

The cordic block 450 receives the I and Q signals of the phase rotatedoutput of the carrier recovery filter 374 and produces a correspondingphase signal 470. The phase-frequency detector 452 then processes thephase signal 470 to produce a phase error signal 472 by comparing thephase signal 470 with a phase reference signal expected for zero phasenoise. The phase reference signal is typically a phase vector that liesalong the I axis.

The phase-frequency detector 452 can operate in different phase trackingmodes including a full phase tracking mode, and an overmodulationhandling mode. The overmodulation handling mode can be a firstovermodulation handling mode or a second overmodulation handling mode.The use of the overmodulation handling mode provides protection againstovermodulation of the picture carrier signal for analog broadcasttelevision signals. For instance, for NTSC television signals, negativemodulation is used and an NTSC television signal has its highest signallevel during the sync interval. However, in some cases, such as whenvery white television components are transmitted, due to overmodulation,the amplitude of the picture carrier signal can tend towards zero, bevery small and less than the noise level, or can even undergo 180degrees phase reversal. It then becomes difficult to track the phase andif one is not careful then noise can be tracked instead of the picturecarrier. When the level of the real picture carrier then becomes largerand is detected, it can be out of phase with the phase that is currentlybeing tracked. This overmodulation situation can be reflected in themagnitude level signal 402, which can be used to obtain accurate resultswithout phase synchronization.

FIG. 12B shows an exemplary signal and defines the percentage of videomodulation. Most specifications define overmodulation as more than 87.5%of modulation as shown in FIG. 12B. However, some television signals aretransmitted with an amount of modulation that is larger than 87.5% andin some cases can even have over 100% modulation. During portions oftelevision signals in which the modulation is greater than 100%, thephase of the picture carrier is reversed.

The phase-frequency detector 452 can operate in the first overmodulationmode or the second overmodulation mode to compensate for overmodulation.Both of these modes are a non-coherent way to detect and handleovermodulation. Both of these modes employ a magnitude level, providedby the magnitude level signal 402, which is the level of the filteredpicture carrier signal to determine when overmodulation has occurred andto gate off unreliable phase information. Accordingly, theovermodulation filter 406 is a lowpass filter in which the cutofffrequency is set to remove rapid transitions and spurious switches inthe picture carrier signal. The phase of the filtered picture carriersignal is not used in either overmodulation mode. Further, the input tothe picture carrier recovery block 366 is used to detect overmodulationrather than the demodulated output (i.e. the output of the VSB filter384).

Referring to FIG. 12C, shown therein is a graphical representation ofthe first overmodulation handling mode. The value of the magnitude levelsignal 402 is compared against a threshold, Noise_thresh, to determineif overmodulation is occurring. When the magnitude level signal 402falls below the threshold, the carrier recovery block 378 enters into aloop frozen mode and essentially holds the phase control signal 404 andthe analog mode frequency shift feedback signal 390 constant by usingthe last or previous phase correction and frequency correction valuesprior to entering the loop frozen mode. This can also be referred to asfree-running phase tracking. Accordingly, in this case, unreliable phaseinformation measured by the phase-frequency detector 452 is preventedfrom being further processed by the phase and frequency correctionstages when the magnitude of the filtered picture carrier signal issmall and unreliable. When the magnitude level signal 402 is greaterthan the threshold, the frequency-phase detector 452 exitsovermodulation handling mode, and the phase control signal 404 and theanalog mode frequency shift feedback signal 390 become active again toactively track the phase and frequency errors. The full value of thephase and frequency errors are used and so it can be considered that aweight having a value of 1 is applied to these errors in this case.

The value of the threshold Noise_thresh can be selected to be the valuethat is expected for the white level of the magnitude level signal 402when overmodulation is at about 90%; i.e. the magnitude levelcorresponding to a white level is 10% for 90% overmodulated video on anormalized basis, and the threshold Noise_thresh can be set to 0.1 interms of normalized magnitude, i.e. the threshold Noise_thresh is at amagnitude level equivalent to a normalized magnitude level of about 10%.This normalization can be done by the magnitude detector 408 or anothersuitable element. This threshold can also be changed depending onwhether the phase and frequency correction stages (i.e. phase andfrequency correction loops) are operating in an acquisition state andare attempting to determine coarse phase and frequency settings, or ifthey are operating in a lock state and have determined the current phaseand frequency operating points. For instance, the threshold Noise_threshcan be set to a lower value during the acquisition state than during thelock state so that the carrier recovery tracks more often during theacquisition state. Alternatively, in other embodiments, the thresholdNoise_thresh can have the same value during both acquisition and lockstates.

In addition, the timing for applying the weights or using a previousphase or frequency correction value with respect to the detection ofovermodulation can be skewed (i.e. advanced or retarded), or stretched.Accordingly, the carrier recovery block 378 can apply a guard band,which is a block of time, that can be skewed or stretched with respectto the onset and termination of overmodulation detection so that duringthe guard band the phase and frequency errors are not being activelytracked. The amount of skewing or stretching can be based on theseverity of the noise at the output of the frequency rotator 364. Thiswill provide robustness under more severe phase noise conditions, suchas under multipath scenarios. In some embodiments, the timing of theguard band for compensating for overmodulation earlier (in terms of thenumber of samples) with respect to the onset of overmodulation and tocontinue compensating for overmodulation for some time afterwards (onceagain in terms of the number of samples) with respect to the terminationof overmodulation can be the same. The number of samples to advance ordelay entry into or exit out of overmodulation compensation with respectto the onset or termination of overmodulation detection can be in therange of 8 to 15 samples. The guard band can also be used in the secondovermodulation handling mode which is now discussed.

Referring to FIG. 12D, shown therein is a graphical representation ofthe second overmodulation handling mode. The second overmodulationhandling mode uses a soft approach to gate off unreliable phaseinformation which would otherwise be measured by the phase/frequencydetector 452 during overmodulation. The second overmodulation handlingmode applies a weight to the phase error signal 472 and the frequencyerror signal 494 before these signals are used in the remainder of thephase and frequency correction stages.

In some embodiments, the weighting is a piece-wise linear function basedon the value of the magnitude level signal 402 as shown in FIG. 12D. Themaximum weight is 1, which is similar to the full phase tracking modewhen the measured phase and frequency errors are used to actively trackphase and frequency in the frequency and phase correction stages. Belowthe threshold Noise_thresh, operation is somewhat similar to the loopfrozen operation for the first overmodulation handling mode in which thephase control signal 404 and the analog mode frequency shift feedbacksignal 390 are held constant by applying a weight of zero to the phaseerror signal 472 and the frequency error signal 494. In this case, theoutput of the filter 454 decays to zero depending on the duration of theloop frozen operation. Generally, a large weight is applied to the phaseerror signal 472 and the frequency error signal 494 when the magnitudelevel signal 402 is large, and a small weight is applied to the phaseerror signal 472 and the frequency error signal 494 when the magnitudelevel signal 402 is small. Operation in the sloped regions can bereferred to as soft phase tracking.

The shape of the piece-wise weighting function can be changed by settingdifferent values for the thresholds Noise_thresh OM1_thresh, andOM2_thresh the first weight, the second weight, and by using differentvalues for the two slopes. In alternative embodiments, more than threethreshold values can be used, and the additional thresholds are used todetermine the amount of soft phase tracking in finer granularities.Also, different values can be selected for these parameters if the phaseand frequency correction stages (i.e. phase and frequency correctionloops) are operating in an acquisition state and are attempting todetermine coarse phase and frequency settings, as compared to when thesestages are operating in a lock state and have determined the currentphase and frequency operating points.

In this exemplary embodiment, the slope of the weighting curve increaseswith magnitude until the maximum weighting is reached. The first weightis set at noise_w, and the threshold Noise_thresh is in the range of amagnitude level that is equivalent to a normalized magnitude level ofabout 5% to 10%. The threshold OM1_thresh is chosen to be greater thanthe threshold Noise_thresh and is generally less than a magnitude levelthat is equivalent to a normalized magnitude level of 0.3. The secondweight is chosen to be greater than the first weight and generally lessthan a magnitude level that is equivalent to a normalized magnitudelevel of 0.2. The threshold OM2_thresh is chosen to be greater than thesecond threshold and is less than a magnitude level that is equivalentto a normalized magnitude level of 1.0. The slopes are chosen so thatthe weighting curve is a piece-wise linear function that connects thedifferent threshold/weighting pairs together. In general, the weightingfunction can be the same during the acquisition and lock states.However, for more phase tracking during the acquisition state, thethresholds can be set lower and the weights can be set higher than thesettings for the thresholds and weights used in the lock state. Timingchanges, similar to those discussed in the first overmodulation handlingmode may or may not be used in the second overmodulation handling mode.

In all of the phase tracking modes, the phase correction loop operateswith the gain value G_(ph) being selected for the phase loop amplifier458 (which can be implemented as a multiplier) to provide a compromisebetween the bandwidth of the phase tracking stage and hence the speedwith which the phase tracking stage can lock onto the phase referencesignal and respond to changes in the phase error signal 472. The amountof filtering provided by the lowpass filter block 454 can also beadjusted to obtain a desired speed and bandwidth for the phase trackingstage. Also, the gain value G_(ph) is selected to provide an acceptableamount of phase error while the phase tracking stage still remains inlock. The value of G_(ph) can also be modified during operationdepending on whether phase or frequency lock has occurred to improve theoperation of the phase tracking stage. As a rule of thumb, the gain andloop bandwidth are selected so that under a step response condition, theresidual phase error does not result in multiple oscillations beforesettling to zero.

In the linear regions of the second overmodulation handling mode, thephase-frequency detector 452 generally operates by multiplying the phaseerror with a positive non-zero weight that is less than 1, so that theeffective gain value G_(ph) of the phase correction loop is set to asmaller value than that used in the full phase tracking mode. Thisallows the phase tracking stage to react more slowly to changes in thephase error signal 472 since the phase error signal 472 is more likelyto be affected by noise due to the lower value of the magnitude levelsignal 402.

In each of the phase tracking modes, the phase error signal 472 isfiltered by the low pass filter 454 to obtain a filtered phase errorsignal 474. The filtered phase error signal 474 is amplified by the loopgain 458 and passed into the phase adjustment block 456. The filteredphase error signal 474 is processed by the phase adjustment block 456which compensates for 180 degrees phase synchronization that may occurin the filtered phase error signal 474. The 180 degrees phasesynchronization may occur during system initialization under very highsignal to noise conditions, where the carrier recovery block 366 canlock onto the 180 degrees operating point. In this exemplary embodiment,the phase adjustment block 456 can also be disabled or enabled.

The phase inversion detector 462 checks the filtered phase error signal474 to determine if there is a +/−180 degrees phase shift in the phaseerror. The phase inversion detector 462 checks for how long the filteredphase error signal 474 is in the region of +/−180 degrees (this can bedone using a counter). If the filtered phase error signal 474 is in theregion of +/−180 degrees for a certain period of time, the phase loop isrunning at an operating point that is off by 180 degrees. The phaseinversion detector 462 then produces a 180 degrees phase inversiondetection signal 476 to indicate whether this 180 degrees phasesynchronization has occurred. The filtered phase error signal 474 isalso amplified by the phase loop amplifier 458 to produce an amplifiedphase error signal 478. The phase inversion block 464 receives both the180 degrees phase inversion detection signal 476 and the amplified phaseerror signal 478. If 180 degrees phase synchronization has not beendetected, then the output of the phase adjustment block 456 is theamplified phase error signal 478; i.e. no correction is applied by thephase inversion block 464. If 180 degrees phase synchronization has beendetected, then the output of the phase adjustment block 456 is a 180degrees phase adjusted signal; i.e. a 180 degree phase correction isprovided by the phase inversion block 464.

The phase oscillator block 460 produces the phase control signal 404based on the output of the phase adjustment block 456. The phaseaccumulator 466 accumulates the values of the amplified phase errorsignal, which has been corrected for 180 degrees phase synchronization,to obtain an integrated phase value. This accumulation, or integration,which can be weighted in some implementations, allows the phase trackingstage to respond a bit slower to instantaneous changes in the phaseerror signal 472 and therefore operate in a more stable fashion. Theintegrated phase value is then converted to I and Q signals by thecordic block 468 which are outputted as the phase control signal 404 andprovided to the phase rotators 376, 382 and 386. When the integratedphase error tends towards zero, the phase tracking stage is locked andthe phase noise in the desired television channel signal is compensated.

The frequency correction stage also includes the cordic block 450, thephase-frequency block 452, a summer 480, a decimation filtering block482, a frequency loop amplifier 484, and a frequency oscillator block486. The frequency oscillator block 486 includes a frequency accumulator488, a frequency clipping block 49 b, a phase accumulator 510, and acordic block 492. The frequency correction stage can also operate in afull frequency tracking mode, and employ soft frequency tracking andfree running frequency tracking, as in the case of the phase correctionstage. The frequency correction stage is a slower loop than the phasecorrection stage. Accordingly, the value for the frequency loop gain−G_(fr) and the bandwidth of the frequency loop is selected so that thestep response for this loop is dampened with little overshoot. Thefrequency correction stage should be able to track within 500 kHz of theactual picture carrier frequency taking into account the frequencyoffset error.

The phase-frequency detector 452 generates a frequency error signal 494.The frequency error signal 494 is derived from the phase error signal472 in that the values in the frequency error signal 494 are a series ofdelta-phase errors since frequency is the derivative of phase. Thefrequency error signal 494 indicates the offset of the picture carrierfrequency from DC at the output of the frequency rotator 364. The phaserotators 376, 382 and 386 can correct for some amount of frequency errorbefore the signal enters into the carrier recovery block 378. As aresult, the amount of frequency correction that is required by thefrequency loop takes into account the amount of frequency error to becorrected by the phase rotators 376, 382 and 386. This information isconsidered by adding the frequency error signal 494 to the output of thephase adjustment block 456 via the summer 480 to produce an adjustedfrequency error signal 496.

The adjusted frequency error signal 496 is then filtered and decimatedby the decimation filtering block 482 to produce a filtered frequencyerror signal 498. The decimation filtering block 482 provides lowpassfiltering to smooth out the values in the adjusted frequency errorsignal 496. Decimation, which is optional, is used in this exemplaryembodiment for increasing implementation efficiency. The filteredfrequency error signal 498 is then amplified by the frequency loopamplifier 484, which can also be implemented as a multiplier, to producean amplified frequency error signal 500. The amount of filteringprovided by the decimation filtering block 482 and the amount ofamplification provided by the frequency loop amplifier 484 can beadjusted to control the bandwidth and hence the speed of the frequencytracking stage.

The amplified frequency error signal 500 is then provided to thefrequency oscillator block 486, which generates the analog modefrequency shift feedback signal 390 based on the amplified frequencyerror signal 500. The amplified frequency error signal 500 is firstprocessed by the frequency accumulator 488, which keeps track of thecurrent frequency and updates it with the current value in the amplifiedfrequency error signal 500 to produce a frequency adjusted signal 502.In some implementations, the frequency accumulator 488 can averageconsecutive values in the amplified frequency error signal 500 prior toadjusting the current frequency value. In some cases, weighted averagingcan be used. The amplified frequency error signal 500 allows forcompensating for the frequency offset error that was discussedpreviously. The frequency accumulator 488 is provided with an initialfrequency value, which is the amount of frequency shift that is expectedto be applied to the processed digitized coarse channel signal 192 tocenter it about DC. At initial operation, the current frequency value isset based on the initial frequency value and thereafter updated based onthe values in the amplified frequency error signal 500.

The frequency clipping block 490 specifies upper and lower limits forthe frequency adjusted signal 502 to define a range of frequencies overwhich frequency tracking operates for picture carrier recovery. Theamplified frequency error signal 500 is added to the current frequencyin the frequency accumulator 488, and the result is compared to amaximum and minimum frequency in the frequency clipping block 490. Ifthe resulting frequency is greater than the maximum or less than theminimum clipper frequencies, the frequency accumulator value is clipped(i.e. limited) to the maximum value limit or the minimum value limit,respectively; hence the feedback signal 512 from the frequency clippingblock 490 to the frequency accumulator 488. In alternative embodiments,the clipping function can be replaced by a wrapping function in which,when one frequency limit is reached without the picture carrier beinglocked, the current frequency used by the frequency accumulator 488 isset to the opposite limit via the feedback connection. The output of thefrequency clipping block 490 is provided to the phase accumulator 510which in turn provides an input to the cordic block 492 which thengenerates the analog mode frequency shift feedback signal 390. The phaseaccumulator 510 keeps track of the current phase. The frequencyaccumulator 488 provides the value by which the phase accumulator 510 isincremented on each cycle after processing by the frequency clippingblock 490. The higher the frequency, the more rapidly the phase willaccumulate.

The status stage includes a lock detector 504. The lock detector 504receives the filtered phase error signal 474 and the filtered frequencyerror signal 498 and determines whether lock has occurred for phase andfrequency tracking. The lock detector 504 provides a phase lock statussignal 506 and a frequency lock status signal 508. These values can bestored in status registers associated with the video processing block182. The values can then be used to modify some parameters in the phaseand frequency tracking stages, as well as some parameters of the blocksin the picture carrier recovery block 366 such as the bandwidth of thecarrier recovery filter 374 and the AGC filter 380. The video processingblock 182 can adopt a different set of parameters during phase orfrequency acquisition versus phase or frequency lock (i.e. forimplementing coarser or tighter search ranges as well as faster orslower response). In at least some embodiments, the lock detector 504can also generate signal B1, which is used to indicate that a lock hasbeen made to the picture carrier, and communicate signal B1 to thecontrol block 190 as explained previously with respect to FIG. 7.

As mentioned previously, various audio standards are used for the audioinformation that is present in analog television signals. For instancein North America, there is only one audio carrier that is used with atelevision signal, however, the audio carrier may carry stereo audioinformation. In Europe, a NICAM standard can be used which is adigitally encoded audio signal that is included with an analogtelevision signal. However, other standards use two analog audio carriersignals to encode “right-sided” audio information and “left-sided” audioinformation. Each of these scenarios can be handled by the first andsecond audio filtering blocks 184 and 186, and the audio processingblock 188. If only one audio carrier signal is used, then just the firstaudio filtering block 184 is enabled. Digital broadcast standardtelevision signals include multiplexed audio and video information. Inthis exemplary embodiment, such television signals are processed by thevideo processing block 182, which provides a digital output containingthe video and audio information as the output signal. The video andaudio information can then be further processed by another element, suchas a downstream digital demodulator (not shown), for example. Theprocessing provided by the first and second audio filtering blocks 184and 186, and the audio processing block 188 can provide SIF (a SoundIntermediate Frequency signal) or baseband sound outputs. With an SIFsignal output, another audio decoder can be connected to the receiver100 to process the audio information.

The structure of the first and second audio filtering blocks 184 and 186are similar. Accordingly, only the first audio filtering block 184 willbe discussed in greater detail with reference to a first embodimentshown in FIG. 13A and an alternative embodiment shown in FIG. 13B. Thesound carrier signal is a narrowband signal that is separated from theaccompanying video information (refer to FIG. 11B for an example). Thefirst audio filtering block 184 is provided with the same processeddigitized coarse channel signal 192 as the video processing block 182 orwith another signal as explained below in relation to FIG. 13B. Audiosignals can have a wide variation in bandwidth; the audio bandwidthrange extends from about 50 kHz to 700 kHz. Accordingly, the first audiofiltering block 184 employs a somewhat similar processing methodology asthe video processing block 182 in that a fixed filter is used along withresampling to make it appear as if the fixed filter has a variablebandwidth. In addition, the first audio filtering block 184 generallyemploys a second frequency tracking loop that is configured to extractthe audio carrier frequency of the desired television channel signal foranalog television broadcast standards and the audio filtering block isconfigured to apply a second known frequency shift to compensate for aknown frequency offset in the audio carrier frequency. However, inalternative embodiments, the frequency tracking in the audio filteringblock can be slaved to the frequency tracking employed by the videoprocessing block 182 as described with relation to FIG. 13B.

Referring now to FIG. 13A, shown therein is an exemplary embodiment ofthe first audio filtering block 186. The first audio filtering block 186includes a frequency rotator 550, a first decimation filtering block552, an audio pre-polyphase filter 554 p, a first audio polyphase filter554, an audio resampling phase control block 556, a second decimationfiltering block 558, a third decimation filtering block 560, amultiplexer 562, an audio filter 564, an audio polyphase filter 566, afrequency demodulator 568 and an audio IF carrier recovery block 570.The audio pre-polyphase filter 554 p and the first audio polyphasefilter 554 can be considered to be an audio polyphase filter stage forresampling a signal at a new sampling rate. The decimation filteringblock 552, audio pre-polyphase filter 554 p, first audio polyphasefilter 554, audio resampling phase control block 556, second decimationfiltering block 558, third decimation filtering block 560, multiplexer562, audio filter 564, and audio polyphase filter 566 can be referred toas an audio filter stage. Also, the second frequency tracking loopincludes the frequency rotator 550, the audio filter stage, thefrequency demodulator 568 and the audio IF carrier recovery block 570.

The frequency rotator 550 receives the processed digitized coarsechannel signal 192 and shifts the frequency content of this signal suchthat the frequency content of the audio of the desired televisionchannel signal is approximately centered about DC. However, due to thefrequency offset uncertainty, exact centering about DC is likely notachieved. The output of the frequency rotator 550 is then filtered anddownsampled by the decimation filtering block 552. The output of thedecimation filtering block 552 is filtered by the audio pre-polyphasefilter 554 p, and subsequently resampled by the audio polyphase filter554 based on a first resampling control signal 572. The audiopre-polyphase filter 554 p is configured and used in a similar manner asthe video pre-polyphase filter 352 p. It should be noted that thefunction of the decimation filtering block 552 can optionally beincluded into the functionality of the audio pre-polyphase filter 554 pand/or the audio polyphase filter 554. The audio resampling controlblock 556 provides the value for the first resampling control signal 572based on the audio broadcast standard that corresponds with thetelevision broadcast transmission standard for the desired televisionchannel signal. The audio polyphase filter 554 then resamples the outputof the audio pre-polyphase filter 554 p so that its bandwidth istransformed to match the bandwidth of the desired audio signal with thebandwidth of the fixed audio filter 564; this operation is analogous tothat in the video processing block 182 and therefore does not need to befurther discussed. In alternative embodiments, the functionality of theaudio resampling control block 556 can be provided by the control block190.

However, since the bandwidth of the desired audio signal ranges from 50kHz to 700 kHz, and a relatively large sampling rate is being used, theaudio filtering block 184 employs the decimation filtering blocks 558and 560 and the multiplexer 562 for more efficient processing. Theseblocks are used to extend the range of bandwidth control by anadditional 2 octaves. The blocks 558 to 562 are not required if theaudio polyphase filter 554 is configured to deal with these differentfrequency ranges for the audio.

In this exemplary embodiment, there are three audio signal pathways fromthe audio polyphase filter 554 to the audio filter 564 via themultiplexer 562. The audio resampling control block 556 provides anaudio pathway selection control signal 574 to the multiplexer 562 toselect one of the three audio pathways. A first audio pathway existsfrom the output of the audio polyphase filter 554 to the audio filter564. A second audio pathway exists from the output of the audiopolyphase filter 554 through the decimation filtering block 558 to theaudio filter 564. A third audio pathway exists from the output of theaudio polyphase filter 554 through the decimation filtering blocks 558and 560 to the audio filter 564.

The first audio pathway does not provide any downsampling, while thesecond audio pathway provides a first amount of downsampling and thethird audio pathway provides a second amount of downsampling that islarger than the first amount of downsampling. Accordingly, the firstaudio pathway can be selected when the desired audio signal has a highbandwidth at the upper end of the audio bandwidth range. The secondaudio pathway can be selected when the desired audio signal has a mediumbandwidth that is somewhere between the lower and upper limits of theaudio bandwidth range. The third audio pathway can be selected when thedesired audio signal has a small bandwidth that is at the lower end ofthe audio bandwidth range. In general, a greater or lesser number ofdecimation stages may be configured depending on the desired range ofaudio bandwidths to be supported.

The audio filter 564 operates in a similar manner as the video filter356 and therefore does not need to be described in detail. The output ofthe audio filter 564 is provided to the audio polyphase filter 566 whichupsamples the audio signal to generate a sound IF (SIF) signal 576. TheSIF signal 576 can be further processed by the audio processing block188. The output of the audio polyphase filter 566 can also be sent tothe frequency demodulator 568 which demodulates this output to producethe intermediate audio signal 196. The frequency demodulator 568 istypically an FM demodulator which is well known to those skilled in theart. If the desired audio signal is a mono audio signal, then theintermediate audio signal 196 is a baseband audio signal. For otheraudio broadcast standards, the intermediate audio signal 196 is anothermodulated audio signal. The sampling rate associated with the SIF signaland the intermediate audio signal 196 can be on the order of 1.536 MHzto improve noise performance.

The audio IF carrier recovery block 570 receives a sound IF carrierrecovery signal 578 from the audio processing block 188 (discussed withrelation to FIG. 14), which is used to track the audio carrier signal.In alternative embodiments, the audio IF carrier recovery block 570 canreceive the output signal of the frequency demodulator 568. However, thesound IF carrier recovery signal 578 is a better quality signal withless noise. The audio IF carrier recovery block 570 tracks an audiocarrier signal that corresponds to the desired television channel signaland provides an audio frequency shift feedback signal 580 to thefrequency rotator 550 for shifting the audio carrier frequency to DCwhen doing baseband demodulation and for shifting frequency content ofthe audio information to DC for SIF only processing. In an alternativeembodiment, the audio IF carrier recovery block 570 is not dependent onthe sound IF carrier recovery signal 578 but rather is configured tooperate in a free running mode such that frequency rotator 550 providesa fixed frequency shift. In this case, audio carrier recovery may beperformed at a later stage without feedback to frequency rotator 550 orto audio IF carrier recovery block 570. The implementation of the audioIF carrier recovery block 570 is known to those skilled in the art.

In an alternative embodiment, the audio filtering blocks 184 and 186have a different configuration as shown by audio filtering block 184′ inFIG. 13B. The input to the audio filtering block 184′ is the output ofthe frequency rotator 350 in the video processing block 182 instead ofthe processed digitized coarse channel signal 192. The operation of theaudio IF carrier recovery block 570 is also modified to compensate forthe frequency shift provided in the output of the frequency rotator 350.Furthermore, in order that the audio signal may benefit from the phaseerror tracking provided by the carrier recovery block 387 of the videoprocessing block 182, the output of the phase accumulator 510 is used toproduce the audio frequency shift signal 580′ which controls thefrequency shift applied by the frequency rotator 550 to its inputsignal: the output of the frequency rotator 350. This can be achieved byadding the output of the phase accumulator 510 in the carrier recoveryblock 378 to the output of a similar phase accumulator (not shown) inthe audio IF carrier recovery block 570. Accordingly, the audio IFcarrier recovery block 570 still generates the free running frequencythat is applied to the frequency rotator 550 in terms of frequency, butwith phase correction provided by the output of the phase accumulator510 to generate the audio frequency shift signal 580′. The end result isthat the output of the frequency rotator 550 is shifted to DC in thesame way that it would have been if the input to the frequency rotator550 had not come from the output of the frequency rotator 350 (i.e. ascurrently described in FIG. 13A). In this way, audio carrier recovery iseffectively slaved to picture carrier recovery. Note that considerableprocessing delay may exist in the video path between the output of thefrequency rotator 350 and the output of the phase accumulator 510.Additional benefit to the reduction of phase noise in the audio path maybe achieved if a similar amount of delay is inserted between the outputof the frequency rotator 350 and the input to the frequency rotator 550.In this way the frequency correction provided by the rotator 350 and thephase correction provided by the phase accumulator 510 are effectivelysynchronized.

Referring now to FIG. 14, shown therein is a block diagram of anexemplary embodiment of the audio processing block 188. The audioprocessing block 188 has a first processing pathway including a firstdecimation filtering block 600, a de-emphasis filter 602, a firstmultiplexer 604, and a first audio polyphase filter 606. The audioprocessing block 188 also includes a second processing pathway includinga second decimation filtering block 608, a de-emphasis filter 610, asecond multiplexer 612 and a second audio polyphase filter 614.

The first and second decimation filtering blocks 600 and 608 each havefirst and second stages that perform decimation filtering. The output ofthe first stage of the first decimation filtering block 600 is connectedto a pilot recovery and audio extraction block 616. The output of thepilot recovery and audio extraction black 616 is connected to the inputof the second stage of the second decimation filtering block 608.

The outputs of the first and second audio polyphase filters 606 and 614are connected to a mixture block 618. The outputs of the first andsecond audio polyphase filters 606 and 614 are also connected tofrequency rotators 620 and 622, respectively, which are both connectedto a summer 624. The output of the mixture block 618 and the summerblock 624 are provided to a third multiplexer 626. The third multiplexer626 is connected to a third audio polyphase filter 628. The audioprocessing block 188 also includes a NICAM processing block 630, a FIFO632 and a NICAM sampling control block 634. The NICAM processing block630 is a combination of a NICAM demodulator and decoder.

In operation, the audio processing block 188 can be provided with avariety of input signals depending on the audio broadcast standard thatis used for providing the audio information of the desired televisionchannel signal. For instance, if one audio carrier is used for thedesired television channel signal, the first decimation filtering block600 is provided with the first audio intermediate audio signal 196.Alternatively, if two audio carriers are used for the desired televisionchannel signal, the first and second decimation filtering blocks 600 and608 are provided with the first and second intermediate audio signals196 and 198 respectively. The first and second de-emphasis filters 602and 610 can also be provided with NICAM data which is further describedbelow.

The first stage of each decimation filtering block 600 and 608 providesa first amount of filtering and downsampling to a first sampling ratesuch that secondary audio program (SAP) and L-R (Left-Right) audioinformation is retained. The SAP and L-R audio information exists fromabout 15 KHz up to about 90 KHz. The second stage of each decimationfiltering block 600 and 608 provides a second amount of filtering anddownsampling to a second sampling rate such that the SAP and L-R audioinformation is removed. The second sampling rate is a quarter of thefirst sampling rate. The output of the second stages of the decimationfiltering blocks 600 and 608 is an FM demodulated audio baseband signal.The output of the second stage of the first decimation filtering stage600 provides the sound IF carrier shift signal 578 to the first audiofiltering block 184. Likewise, the output of the second stage of thefirst decimation filtering stage 608 provides a sound IF carrier shiftsignal 578′ to the second audio filtering block 186.

Any DC level in the sound IF carrier shift signal 578, which is an FMdemodulated audio baseband signal, indicates that the mixing frequencyof the frequency rotator 550 isn't exactly set to the carrier frequency.This can be understood since FM modulation changes the carrier frequencybased on the instantaneous level of the modulating signal. If themodulating signal (at the transmitter) was at a DC level, this would beindistinguishable from a non-modulated carrier signal at a slightlydifferent frequency. Accordingly, any DC that is present in the sound IFcarrier shift signal 578 must be due to the mixing frequency of thefrequency rotator 550 being different from the actual transmittedcarrier frequency and the audio carrier IF recovery block 570 accountsfor this difference in the audio frequency shift feedback signal 580.

The pilot recovery and audio extraction block 616 receives the SAP andL-R audio information from the first stage of the decimation filteringblock 600 and locks to the pilot tone to demodulate the L-R audioinformation. The pilot recovery and audio extraction block 616 can alsoprovide a locked carrier for SAP demodulation. The demodulated L-R audioinformation (i.e. BTSC L-R audio signal 638) enters the second stage ofthe decimation filtering block 608, and gets filtered and downsampled to48 KHz, with content up to only about 15 KHz. Those skilled in the artare familiar with the implementation of the pilot recovery and audioextraction block 616.

SAP and L-R audio information are present in North American BTSC signals(this audio standard somewhat corresponds to the NTSC video standard).Broadcasters in North America don't have to transmit SAP or stereoinformation. With other standards in other countries, the FM signal maycontain other information, or have that information in another format.For example, in Japan, the L-R audio information sits in the FMdemodulated signal just above the L+R audio information, similar to theBTSC signal, but is FM modulated instead of AM modulated. Some standardsalso transmit a “mode tone” within the FM demodulated signal, just abovethe audio portion. This can be used to indicate if a stereo signal ispresent, or that perhaps a second language is being transmitted.

FM audio signals are broadcast with pre-emphasis which boosts the highfrequency content of the FM audio signals above high frequency noisewhich is inherent in FM transmission. Accordingly, the de-emphasisfilters 602 and 610 are used to perform the opposite operation,de-emphasis or an attenuation of high frequencies, to restore thefrequency content of the FM audio signal to its original levels. The L-Raudio information is broadcast using a more complicated pre-emphasisfunction, requiring a corresponding de-emphasis function (wDBX) thatremoves the pre-emphasis. The de-emphasis functions performed by thede-emphasis filters 602 and 610 are known to those skilled in the art.

The output of the pilot recovery and audio extraction block 616 canprovide SAP and EIAJ (the Japanese standard) L-R audio information. TheEIAJ audio signal is FM modulated within the FM demodulated spectrum,above the baseband audio information. The second audio filtering block186 can be used to FM demodulate this signal. For example, in EIAJ, thefirst audio filtering block 184 demodulates the broadcast FM signalwhich contains audio information from 0-15 KHz, FM modulated audioinformation centered at ˜31 KHz, and additional audio information. Thissignal gets routed to the second audio filtering block 186 to beFM-demodulated in order to extract the additional audio information.

The inputs of the multiplexer 604 are the output of the de-emphasisfilter 602 and the SIF1 signal provided by the first audio filteringblock 184. Likewise, the inputs of the multiplexer 612 are the output ofthe de-emphasis filter 610 and the SIF2 signal provided by the secondaudio filtering block 186. Either the SIF1 and SIF2 signals are selectedas the outputs of the multiplexers 604 and 612 or the outputs of thede-emphasis filters 602 and 610 are selected according to a selectioncontrol input 640 that can be pre-defined or user defined.

The outputs of the multiplexers 604 and 612 are then provided to theaudio polyphase filters 606 and 614 which restore the sampling rate to apower of 2 division of the clock rate of the ADC 106. However, whenNICAM audio signals accompany the desired television channel signal, there-sampling rate of the polyphase filters 606 and 614 is selected in adifferent manner as is described in further detail below.

The outputs of the audio polyphase filters 606 and 614 are provided tothe mixture block 618. The mixture block 618 merges the 2 audio channelsignals appropriately to generate a 2 channel output (i.e. a left andright output).

Alternatively, when the audio processing block 188 provides an SIF audiooutput, the SIF1 and SIF2 audio signals are selected by the multiplexers604 and 612 respectively, resampled by the audio polyphase filters 606and 614 respectively and provided to frequency rotators 620 and 622. TheSIF1 and SIF2 audio signals are then shifted in frequency according tothe IF frequency shift signals 642 and 644, and are then summed by thesummer 624. Both frequency rotators 620 and 622 are enabled when twosound carriers are broadcast with the desired television channel signal.

The multiplexer 626 selects between the output of the mixture block 618and the output of the summer block 624 based on a selection controlsignal 646 which can be pre-defined or user-programmed depending on themode of operation of the receiver 100 which dictates the type of audioinformation that it should be providing. Alternatively, in at least somecases, the control signals can be provided by the control block 190 whenthe broadcast transmission standard has been detected. The multiplexer626 selects the output of the mixture block 618 when the audioprocessing block 188 is configured to output an audio baseband signal.The multiplexer 626 selects the output of the summer 624 when the audioprocessing block 188 is configured to output an SIF audio signal. Inboth cases, the output of the multiplexer 626 is provided to the audiopolyphase filter 628, which resamples the audio information tocorrespond with the clock rate of the ADC 106 and produces the outputaudio signal 200. Accordingly, the output of the audio polyphase filter628 is either a baseband audio signal, which can be mono or stereodetermined by the mixture block 618, or an SIF signal which can consistof up to two FM carriers, at some programmable frequencies.

NICAM is a digital audio transmission standard. The NICAM processingblock 630 processes one of the SIF1 and SIF2 signals and locks to thesymbol rate and extracts and decodes the transmitted digital data toproduce the decoded NICAM audio signal 636. Those skilled in the art arefamiliar with NICAM demodulation and decoding. However, the symboltiming/period for the symbol rate is defined at the transmitter, whichsent the desired television channel signal, and is unrelated to thesampling rates used in the receiver 100. Nevertheless, the decoded NICAMaudio signal 636 must be correctly output by the DAC 110, which isasynchronous to the transmitter, at the same rate.

The correct output rate can be determined by the NICAM sampling controlblock 634 via the FIFO 632. The output of the FIFO 632 is then providedto the de-emphasis filters 602 and 610. The FIFO 632 is a data structureand in alternative embodiments can be replaced with a memory elementsuch as the on-chip memory (not shown) of the digital processing block108. The correct output rate can be determined by observing the“fullness” of the FIFO 632. For instance, if the FIFO 632 is more thanhalf-full with decoded NICAM audio data, then the output rate of thedecoded NICAM audio data is too slow. In this case, the NICAM samplingcontrol block 634 can increase the sample rate that is applied by theaudio polyphase filters 606 and 614 to output the decoded NICAM audiodata at a faster rate. Alternatively, if the FIFO 632 is less thanhalf-full with decoded NICAM audio data, then the output rate of thedecoded NICAM audio data is too fast. In this case, the NICAM samplingcontrol block 634 can decrease the sample rate applied by the audiopolyphase filters 606 and 614 to output the decoded NICAM audio data ata slower rate.

To initialize the operation of the NICAM processing, the length of timerequired for the FIFO 632 to reach “half-fullness” is measured. Thistime is then used to set the initial, nominal rate of resampling that isused in the audio polyphase filter blocks 606 and 614. The measurementof the “half-fullness” time and setting of the resampling rate isperformed by the NICAM sampling control block 634. In other words, therate at which the NICAM data is produced is measured so that the rate atwhich this data needs to be output can be determined. With respect toNICAM processing, the audio polyphase filter 628 has a fixed upsamplingrate, and the adjustable audio polyphase filters 606 and 614 areconfigured to have a fixed output rate. Accordingly, changing theresampling rate of the audio polyphase filters 606 and 614 changes onlythe rate of their consumption of input data from the FIFO 632 in orderto maintain the “half-fullness” of the FIFO 632.

It should be noted that in alternative embodiments of the receiver 100,depending on the type of output signal 112 that is desired, such as justa single SIF output signal, only one input audio filtering block isneeded as well as only a portion of the audio processing block 188.Specifically, if the SIF output is all that is needed, then only themultiplexer 604 (with the SIF1 always selected as the input), audiopolyphase filter 606, frequency rotator 620, summer 624 (though now theinput from the frequency rotator 622 is not present so the summer 624becomes a simple pass-through block), the multiplexer 626 (with theoutput of the summer 624 always being the selected input) and the audiopolyphase filter 628 are required. In such embodiments, the sound IFcarrier shift signal 578 will not be provided. In this case, the audiocarrier recovery can be a slave to the video carrier recovery that isperformed in the video processing block 182, as explained previously, sothat once a lock is made to the video carrier signal, a similar lock canbe made for the picture carrier signal. Alternatively, the first audiofiltering block 184 can operate in a free running mode in which case thefrequency relationship between the picture carrier signal and the audiocarrier signal, which is defined by the television broadcast standardfor the desired television channel signal, is used to determineappropriate values for the audio frequency shift feedback signal 580once a lock has been made for the video carrier signal.

The universal television receiver 100 implements a two-stage gaincontrol technique that provides variable gain in both the analog domain(i.e. in the RF and analog processing blocks 102 and 104 (as describedbelow)) and the digital domain (i.e. in the video processing block 108and the audio filtering blocks 184 and 186). The analog gain controlblock and the digital gain control block of the receiver 100 can beconsidered to be the components of a gain control system that is used tocontrol the level of analog and digital gain amplification.Conventionally, variable gain is applied only in the RF and analogprocessing blocks 102 and 104. However, the universal televisionreceiver 100 provides variable gain in both the analog and digitaldomains to provide another level of flexibility in gain control thatresults in improved signal quality in the desired television channelsignal 112.

The analog gain control block 308 provides at least one analog gaincontrol signal to control an amount of analog amplification applied byat least one analog VGA in the receiver. A digital gain control blockprovides at least one digital gain control signal to control an amountof digital amplification applied by at least one digital VGA in thereceiver. The digital gain control block can set a gain coefficient forat least one digital VGA based on a metric of the desired televisionchannel signal. The metric can be one of Signal to Noise ratio, Signalto Noise plus distortion ratio and Bit Error Rate. The metric isselected in part depending on whether the desired television channelsignal is transmitted according to an analog or digital broadcaststandard. The digital gain control block can be block 362 or a gaincontrol block in a digital demodulator (see FIG. 17). In someembodiments, the analog gain control block 308 is operable in first andsecond modes. In the first mode, the analog gain control block 308generates a quasi peak measurement of a digitized version of the desiredtelevision channel signal and utilizes the quasi peak measurement in afeedback loop to control the amplification of at least one analog VGA.In the second mode, the analog gain control block 308 is configured toset an initial gain coefficient of at least one analog VGA based on ametric of the desired television channel signal. In alternativeembodiments, the analog gain control block 308 only operates in thefirst mode or the second mode.

In the analog domain, the gain control signals provided to the variousVGAs can be controlled so that the gain is distributed between thevarious VGAs in a more effective manner. This can be done using avariety of techniques in the first and second modes. The first modeemploys a technique based on recognizing the differences in analog anddigital television broadcast standards and accounting for thesedifferences when determining an effective analog gain distribution sothat the input range of the ADC 106 is effectively utilized in bothcases. This technique is described in further detail with regards toFIGS. 16A-16C. The second mode employs a technique that uses signal andnoise information, and digital and analog metrics to take distortioninto account. This technique is described in further detail with regardsto FIG. 15. In contrast, conventional techniques control two or moreanalog VGAs by simply using measured output levels in which the outputscontain more than just the desired television channel signal and nottaking into account differences in analog and digital televisionbroadcast standards.

In the digital domain, digital variable gain amplification is used toensure an adequate signal level for the desired television channelsignal after filtering by the video filter 356. The amount of filteringperformed by the video filter 356 is known a priori but the levels ofthe interferers will vary which will affect the signal level of thedesired television channel signal. Accordingly, the digital VGA 358 canapply gain to increase the level of the desired television channelsignal when operating in the analog operation mode. In embodiments inwhich the receiver 100 is connected to a downstream digital demodulator(not shown), the demodulator can adjust the digital gain control signal400 for appropriate amplification when the video processing block 182 isoperating in digital operation mode. Although not shown, a digital VGAsimilar to the digital VGA 358 can be connected between the audio filter564 and the audio polyphase filter 566 in order to compensate for thereduction in audio level due to filtering and the level of interferersthat are present. In this case, audio gain control can be eitherindependently controlled based on a desired audio level or can be slavedto the digital gain control signal 398.

In an exemplary embodiment, the gain control method used herein does notrely solely on signal levels, but employs performance criteria for thedemodulated video signals to more effectively set the gain settings atvarious locations in the RF processing block 102, the analog processingblock 104 and the digital processing block 108. The performance criteriathat are used can be the Bit-Error Rate (BER) for digital broadcasttelevision signals, and Signal-to-Noise Ratio (SNR) orSignal-to-Noise+Distortion Ratio (SNDR) for analog broadcast televisionsignals. Conventional gain control schemes only look at the signalstrength right at the output of a variable gain amplifier and apply anamount of gain commensurate with the degree to which the signal strengthis below some established level. However, it is important to note thatthe signal at the output of the variable gain amplifier can include morethan the signal of interest and hence the measured level is not a truemeasure of the signal level. Accordingly, one of the gain controlmethods described herein controls the gain of various variable gainamplifiers by determining the signal quality of the demodulated videosignal and in at least some cases can allow some level of distortion tooccur assuming that the signal artifacts introduced from the distortionproducts do not affect the desired television channel signal more thanan allowable and measurable amount.

Referring now to FIG. 15, shown therein is a flow chart diagram of anexemplary embodiment of a gain control method 650 that can be employedby the universal television receiver 100 to determine the settings forthe various VGAs in the RF and analog processing blocks 102 and 104. Thegain control method 650 involves performing a calibration measurementwhen the universal television receiver 100 is first used, and thenrepeating calibration thereafter from time to time to account for anychanges in the environment. For example, calibration can be carried outon power-up or when changing channels. The gain control method 650 canalso account for temporary interference such as planes flying close by,for example.

The gain control method 650 starts at step 652 in which a desiredtelevision channel is selected. The method 650 then moves to step 654 atwhich point a first combination of gains is applied to the VGAs in theRF and analog processing blocks 102 and 104. At step 656, the quality ofthe demodulated desired television channel signal is measured using aperformance metric. The performance metric can be an analog metric suchas SNR or SNDR to provide information on signal amplitude and signaldistortion when the desired television channel signal is broadcastaccording to an analog standard For example, the SNDR metric can be usedto measure distortion. Alternatively, the performance metric can be adigital metric such as BER when the desired television channel signal isbroadcast according to a digital standard. The signal quality (SNR,SNDR, BER or the like) is measured in the digital processing block 108and adjustments are made to the gain control signals used to control thegain of the various analog and digital variable gain amplifiers toimprove this metric. When varying the gain of the VGAs at step 654,changes to the gain coefficients are made while being careful not tooverload the ADC 106. For instance, if in step 654 the gain is increasedfor a VGA in an earlier stage, then a proportional reduction may beneeded for a VGA at a later stage to avoid overloading the ADC 106.

At step 658, the method 650 determines whether measurements have beenmade for a desired set of gain value (i.e. gain settings or gaincoefficients) combinations of the various VGAs. If not, the method 650goes to step 654 to apply another combination of gain values to the VGAsin the RF and analog processing blocks 102 and 104 and the digital VGA358. If all of the desired gain value combinations have been tried, themethod 650 then goes to step 660 in which the best combination of gainsettings for the current television channel is saved in a look-up tablein the memory of the digital processing block 108. The best combinationof gain settings is selected such that the input range of the ADC 106 iseffectively utilized and there is an acceptable level of signal qualityin the demodulated desired television channel signal determined by theperformance metrics and acceptable signal quality criteria (this can beobtained from the television broadcast standards). These initial gaincoefficient settings for the analog VGAs can then be stored for a giventelevision channel in a gain coefficient table. The table can be indexedaccording to television channel signal and during operation the gaincoefficient settings can be selected from the gain coefficient look-uptable based on the desired television channel signal. The gaincoefficient table is essentially a look-up table. At step 662, themethod 650 determines whether the gain settings for other televisionchannels must be calibrated. If so, the method 650 goes to step 652; ifnot, the method 650 goes to step 664.

In an example implementation, steps 654 to 660 can involve determining afirst gain coefficient for at least one VGA for a nominal desired powervalue for the coarse channel signal 162, measuring a metric for thedesired television channel signal, repeating the setting and measuringsteps for several different gain coefficients and desired power valuesabove and below the nominal desired power value; and selecting the gaincoefficient providing the best value for the metric.

Alternatively, instead of keeping track of the measured metrics for allgain combinations and then selecting the gain coefficients that led tothe best metric, the best gain coefficients can be tracked ascalibration is performed by observing if the metric decreases after again coefficient change; if a decrease occurs then the gaincoefficient(s) can be reverted one iteration. This method of gaincoefficient selection can be carried out for the gain coefficients forall VGAs or first for the gain coefficients of the VGAs in the RF stage(i.e. RF processing block) and then for the gain coefficients of theVGAs in the IF stage (i.e. analog processing block) until completed.

The source of the distortion due to a particular processing block isnever known, thus one typically performs some iterative adjustmentsusing this technique. Accordingly, at steps 654 to 658, another approachcan be to iterate through various gain coefficient settings for thevarious analog and digital VGAs and determine the maximum gain without areduction in the measured metric. The gain coefficient for each VGA canthen be adjusted upwards and downwards to determine the impact on theanalog or digital metric, as the case may be, by introducing intentionaldistortion in order to determine how high the gain control signals canbe set such that the distortion does not appreciably affect the qualityof the desired television channel signal after demodulation and thesignal quality measured by the analog or digital metrics increase.

In another alternative embodiment, the current gain coefficient isobtained by measuring a first gain coefficient of a VGA to achieve adesired power value for the coarse channel signal 162, measuring asecond gain coefficient of that VGA to achieve a desired value for themetric of the desired television channel signal, and calculating adifference gain coefficient or offset from the difference of the firstand second gain coefficients. This VGA is now calibrated. Aftercalibration, during use, the power at the output of the calibrated VGAcan be measured, a third gain coefficient can then be calculated toachieve the desired power value and the third gain coefficient can thenbe adjusted by the difference gain coefficient for the calibrated VGA.The adjusted third gain coefficient is then used as the gain coefficientfor the VGA. This calibration process can be performed for more than oneVGA.

The various gain coefficient selection methods can also be performed forany digital VGAs that are used so that the gain coefficient look-uptable can include gain coefficient settings for analog and digital VGAs.As before, the table is indexed according to television channel signaland during operation the gain coefficient settings for the analog anddigital VGAs can be selected based on the desired television channelsignal that is selected.

At step 664, the gain settings have been calibrated for all televisionchannels. At this point, the method 650 monitors whether there are anytemporary interferers, such as an airplane that flies close by, forexample. If not, the method 650 moves to step 668. One technique fordetermining interference involves determining an expected power levelfor the desired television channel signal and then monitoring the powerlevel of the desired television channel signal during use for anyvariations from the expected power level. If a temporary interferer isdetected at step 664, the method 650 moves to step 666 to perform anadjustment with the analog gain control settings to compensate for thetemporary interference. For instance, gain adjustment may be made for asingle VGA, such as the VGA 158, to deal with a temporary interferer.The VGA 158 can be a “fine” analog VGA that has a fine step size for itsgain settings while the other analog VGAs can be provided with coarsestep size for their gain settings. Alternatively, more than one VGA canhave fine setting control. Alternatively, one or more VGAs can be usedthat have a gain coefficient that is continuously variable. For atemporary interferer, the gain of the VGA 158 can be adjusted tominimize the influence of the interferer on the processed televisionchannel signal 112. When the temporary interference is gone, the gainsetting of the VGA 158 can be set based on the look-up table and themethod 650 moves to step 668. If altering the gain coefficient for thesingle VGA does not provide sufficient gain variation to compensate forthe detected temporary interference, then the analog gain control blockadjusts the gain coefficient of at least another one of the VGAs tocompensate for the detected temporary interference. The adjustment ofthe gain coefficients for the other VGAs can be made one at a time, thatis for the first additional VGA, if adjusting the gain coefficient doesnot provide sufficient gain variation to compensate for the interfererthen the gain coefficient for another VGA can be adjusted and so on andso forth.

At step 668, the method 650 determines whether it is time to performanother calibration. If so, the method 650 goes to step 652. If not, themethod goes to step 664 and monitors for any temporary interferers. Thiscalibration can be done in a periodic manner.

Accordingly, the technique of using digital and analog metrics to setthe gain of the digital and analog variable gain amplifiers is used todetermine the initial point at which the gain coefficients are set forthe various VGAs. In this case, the initial gain coefficients areselected from the perspective of the quality of the demodulated desiredtelevision channel signal rather than simply relying on power levels atthe output of a VGA as is done conventionally. These gain coefficientscan be used until calibration is next performed. Alternatively, thecurrent gain coefficient for a VGA can be generated by incrementing ordecrementing a previous gain coefficient based on a current measuredpower level of the desired television channel signal. Immediatelyfollowing calibration, the previous gain coefficient is the initial gaincoefficient obtained from calibration.

As previously mentioned, the purpose of the analog gain control block308 is to adjust the gain settings at the RF and IF stages to adjust thelevel of the signal presented to the ADC 106 in order to improve thequality of the desired television channel signal. In an alternative, inaccordance with the first mode mentioned previously, this can be doneusing an analog gain control feedback loop in which the output level ofthe ADC 106 is measured and compared to preset reference levels for bothanalog and digital broadcast transmission standards. If the measuredlevel is less than the reference level then one or more gain controlsignals are increased, while if the measured level is higher than thereference level then one or more gain control signals are decreased.Therefore, the reference level may be thought of as a target level whichthe analog gain control feedback loop seeks to maintain, even when thelevel of the signal from the antenna 120, or other input means as thecase may be, changes or the broadcast transmission standard changes forthe desired television channel signal.

However, the characteristics of television signals transmitted underanalog and digital broadcast standards, referred to herein as analog anddigital broadcast television signals respectively, are fundamentallydifferent from one another and as such the best level for digitizationis different in each case. A normally (negatively) modulated analogbroadcast television signal is at its highest level during thetransmission of synchronizing pulses (see FIG. 12B for example). Duringthese periods, the picture carrier signal resembles a pure sinusoidwhich has a peak to average power ratio of 3 dB. Such a signal can beoptimally digitized by the ADC 106 by adjusting the amplitude of thesignal such that its peaks are not clipped and a certain amount ofminimum headroom is maintained with respect to the full-scale range ofthe ADC 106. In practice, several dB of headroom should be provided toallow for measurement error and dynamic effects. Conversely, digitalbroadcast signals may have peak to average power ratios of 10-15 dB ormore. When digitizing these types of signals, the amplitude of thesesignals should be adjusted so that excessive clipping does not occur,though some amount of clipping may be acceptable. If the output levelfrom the ADC 106 were to be calculated based only on the average powerlevel, then the reference level could be set to be optimal fordigitizing analog broadcast television signals or for digitizing digitalbroadcast television signals but not for both. Furthermore, since thecoarse channel signal 162 presented to the ADC 106 may simultaneouslycontain both analog broadcast and digital broadcast television signals,the use of average power as a measure of level is not optimal.Similarly, the use of peak power as a measurement is also not optimal.

Referring now to FIG. 16A, shown therein is a block diagram of anexemplary embodiment of the analog gain control block 700, which employsa quasi peak detector to provide a measure of signal level for theoutput of the ADC 106. The signal level measure is used to provide moreeffective level control when digitizing both analog and digitalbroadcast television signals. Furthermore, the level control iseffective regardless of whether the input to the ADC 106 is dominated bythe desired television channel signal or an adjacent television channelsignal and regardless of whether the dominant television channel signalis transmitted using analog or digital broadcast standards. The analoggain control block 700 generally tries to maintain a 7-10 dB back-off inRMS level during synchronizing intervals for analog broadcast televisionsignals, 12 to 15 dB back-off in RMS level for digital broadcasttelevision signals, and somewhere in between when the digitized coarsechannel signal 172 from the ADC 106 includes both signal typessimultaneously. The back-off is measured relative to the full scalerange of the ADC 106. Also, the analog gain control block 700 attemptsto split the calculated gain between variable gain amplifiers in the RFprocessing block 102 and the IF variable gain amplifiers in the analogprocessing block 104.

The analog gain control block 700 is configured to generate a quasi peakmeasurement to track a level substantially equal to a mean-square levelbased on the digitized coarse channel signal 172 during synchronizingintervals when the desired television channel signal is transmittedaccording to an analog broadcast standard and to track a levelsubstantially greater than the mean-square level based on the digitizedcoarse channel signal 172 when the desired television channel signal istransmitted according to a digital broadcast standard. When the desiredtelevision channel signal is transmitted according to an analogbroadcast standard, the reference level is selected to provide a firstamount of headroom between a root-mean-square level of the digitizedcoarse channel signal and the full-scale range of the ADC 106 duringsynchronizing intervals. When the desired television channel signal istransmitted according to a digital broadcast standard, the selectedreference level provides a second amount of headroom between theroot-mean-square level of the digitized coarse channel signal 172 andthe full-scale range of the ADC 106. The second amount is greater thanthe first amount.

The analog gain control block 700 includes a Power Detector (PD) 702, alowpass filter 704 and a leaky peak detector 706 that together provide arobust measurement of the output level of the ADC 106. The analog gaincontrol block 700 also includes a comparator 708, a decimation block710, a low pass filter 712, a switch 714, an IF gain adjustment path andan RF gain adjustment path. The IF gain adjustment path includes amultiplier 716, a summer 718, an accumulator 720 and a DAC 722. The RFgain adjustment path includes a multiplier 724, a summer 726, anaccumulator 728 and a DAC 730. The analog gain control block 700 canalso include an instability monitor 732 that is optional depending onthe implementation of the ADC 106. The gain adjustment path from theinstability monitor 732 can also be turned off because the bandwidth ofthe lowpass filter 704 can be chosen such that the measurement signalfrom the leaky peak detector 706 is large when the ADC 106 is unstableand in this case the analog gain control block 700 is configured toreduce the analog gain control signals that it provides to bring the ADC106 back to a stable state. However, the instability monitor 732 can beused to detect and/or reset the ADC 106 when it is unstable.

The switch 714 and the RF gain adjustment path can also be optional ifthe analog gain control block 700 is used with a receiver that has an RFprocessing block that is not capable of receiving a gain control signalor provides its own gain control.

The power detector 702 receives the digitized coarse channel signal 172from the ADC 106 and determines the power of this signal by squaring themagnitude of the real and imaginary components of this signal. The powerdetector 702 provides a power signal that is filtered by the lowpassfilter 704. The lowpass filter 704 can be a first order wide-band IIRfilter. The filtered power signal is then processed by the leaky peakdetector 706 which tracks the peak of the output of the lowpass filter704 and outputs a measurement signal. Since, the input signal isfiltered by the lowpass filter 704 before being provided to the leakypeak detector 706, the leaky peak detector 706 will track quasi peaks,the amplitude of which depends on the amount of lowpass filtering andthe nature of the input signal, but the quasi peak is lower than theactual peak of the input signal over a period of time. Accordingly, theoutput of the leaky peak detector 706 can be configured to track theaverage power (mean-square voltage) of the digitized coarse channelsignal 172 during synchronizing intervals for analog broadcaststandards. The output of the leaky peak detector 706 will track a levelgreater than the average power of the digitized coarse channel signal172 for digital broadcast standards.

Referring now to FIG. 16B, shown therein is a block diagram showing thefunctionality for an exemplary embodiment of the leaky peak detector706. The leaky peak detector 706 tracks the peak of its input and“leaks” over time. The leaky peak detector 706 can be implemented, froma functional point of view, using a comparator 732, a multiplier 734, asummer 736, a register 738, a switch 740, a subtractor 742, a register744, a multiplier 746 and a switch 748. The leaky peak detector 706 alsoemploys several parameters: a constant small decay parameter, a fastdecay parameter and an attack parameter. It should be noted that theleaky peak detector 706 may be implemented using dedicated hardware orvia computer code that implements the functionality of the blocks shownin FIG. 16B and is executed by a DSP.

The comparator 732 compares the level of the input signal to the leakypeak detector 706 with the previous peak value that is stored in theregister 738. If the input level is larger than the previous peak value,then the difference between the input level and the previous peak valueis multiplied by an attack parameter via the multiplier 734, added tothe previous peak value by the summer 736 and provided by the switch 740as the output of the leaky peak detector 706. The switch 740 performsthis function when notified by the comparator 732 that the input levelis greater than the previous peak value. This current peak value is alsostored in the register 738 as the previous peak for the next operationof the comparator 732. The scaling of the output of the comparator 732by the attack parameter determines how fast the leaky peak detector 706reacts to peaks in the input signal.

On the other hand, when the input level is smaller than the previouspeak value, the peak output value is decremented by a current decayparameter. In this case, the current decay parameter is subtracted fromthe previous peak value by the subtractor 742 and this reduced peakvalue is provided to the switch 740 which provides this reduced peakvalue as the output of the leaky peak detector 706. Also, the previouspeak value is updated with this reduced peak value (i.e. decrementedprevious peak value). The switch 748 selects either a constant smalldecay value or a multiplied version of a fast decay value as the currentdecay parameter that is stored in the register 744. The fast decay valueis multiplied by the current decay parameter stored in the register 744.The switch 748 selects the multiplied version of the fast decay valuewhen the input signal level is less than the previous peak value for acertain period of time, otherwise the constant small decay value isselected by the switch 748.

In general, the value of the attack parameter is chosen such that theleaky peak detector 706 is reasonably responsive to peaks but notover-reactive to noise. The value of the decay parameter is chosen to beconstant to avoid fluctuations due to analog broadcast video content.Since the decay parameter is small under normal operation, the leakypeak detector 706 will take a long time to decay and catch the peak ofthe input signal in the event of a significant drop in the amplitude ofthe input signal. Accordingly, a general rule for the parameters is tohave fast attack and slow decay constants because the operation of theleaky peak detector 706 should be content independent (i.e. the outputof the leaky peak detector 706 does not vary too much during activevideo lines). To improve performance, the leaky peak detector 706 isconfigured to enter into a fast-decay mode when it does not find a peakwithin a certain period of time. In the fast-decay mode, the value ofthe fast decay parameter is scaled recursively by the multiplier 746until the peak output signal differs from the input signal level by acertain amount at which point the current decay value stored in theregister 744 is updated with the value of the constant small decayparameter.

For example, for operation at 288 MHz, the decay parameter can vary invalue from 0.0000004768 to 1. In at least some embodiments, the value ofthe decay parameter can vary from 0.0000004768 to 0.000005. In at leastsome embodiments, a value of 0.0000008 can be chosen for the decayparameter. The decay parameter should have a small value because theoutput of the leaky peak detector 706 should not fluctuate too much. Thefast decay parameter can in general vary in value from 1 to 4096. In atleast some embodiments, the value of the fast decay parameter can varyfrom 2 to 512. In at least some embodiments, for legacy (i.e. thedesired television channel signal is transmitted according to an analogbroadcast standard) negative modulated signals a value of 256 can bechosen for fast step response. However, for legacy positive modulatedsignals, since the video content is above the sync tip, the fast decayparameter should have a small value, such as 2, to avoid a sudden dropin the peak value due to television signal content variation. For adesired television channel signal that is transmitted according to adigital broadcast standard, a value of 256 can be chosen for the fastdecay parameter for a fast step response. The attack parameter can ingeneral vary in value from 0.000000476 to 1. In at least someembodiments, the value of the attack parameter can vary from 0.00003 to0.02. The actual value is selected based on the input signal statistic.In at least some embodiments, for legacy signals, the leaky peakdetector 706 tracks the peak so a value of 0.002 can be selected for theattack parameter. For a desired television channel signal that istransmitted according to a digital broadcast standard, the leaky peakdetector 706 tracks closer to the RMS value so a value of 0.0002 can bechosen for the attack parameter. It should be noted that the range ofthese parameters provided herein is defined by the number of bits (i.e.precision) in the digital circuit design.

Referring once again to FIG. 16A, the comparator 708 then produces again error signal by comparing the measurement signal with a referencelevel. The reference level is a target level that the analog gaincontrol block 700 attempts to track for the output level of the ADC 106.Since the ADC 106 operates at a high frequency, such as 288 MHz forexample, the gain error signal is greatly oversampled. Furthermore,since the AGC gain control block 700 does not have to control the IFgain and RF gain at such a high rate, the gain error signal is decimatedby the decimation block 710 to a much lower rate on the order of severalhundred kHz. This also reduces the bit precision required for thelowpass filter 712.

The reference level is selected such that during normal closed loopoperation for analog broadcast television signals, the gain of the RFand IF VGAs will be adjusted to provide about 7-10 dB of back-offbetween the RMS level of the input signal to the ADC 106 duringsynchronizing intervals and the full scale range of the ADC 106. Sincethe leaky peak detector 706 tracks a level greater than the averagepower for digital broadcast television signals, the amount of back-offbetween the RMS level of the input signal to the ADC 106 and the fullscale range of the ADC 106 will be larger for digital broadcasttelevision signals than analog broadcast television signals. Thisdifference in the amount of “back-off” will be equal to the differencein tracking level between the analog and digital broadcast signals.Depending on the configuration of the low pass filter 704 and the leakypeak detector 706, the difference may be on the order 5 dB resulting ina total back-off of 12-15 dB for digital broadcast television signals.In this way, the back-off is automatically adapted for effectivedigitization of the type of signal being received, without a prioriknowledge of whether an analog or digital broadcast is being received.This is also effective when the coarse channel signal 162 may containboth signal types and where the relative power between them is unknown.

The gain error signal is then filtered by the lowpass filter 712 toproduce a filtered gain error signal. However, in embodiments whichinclude the instability monitor 732, the analog gain control block 700is configured to select between the gain error signal from thecomparator 708 and a gain adjustment signal provided by the instabilitymonitor 732 as the signal which is filtered by the lowpass filter 712.In either case, the IF and RF analog gain control signals will changeaccordingly. In addition, the IF and RF analog gain control signals canbe maintained at a previously calculated value, if this is requiredduring operation. The operation of the instability monitor 732 isdiscussed in further detail below.

The filtered gain error signal is then scaled by an IF loop gain andaccumulated in the IF gain adjustment path or scaled by an RF loop gainand accumulated in the RF gain adjustment path depending on theoperation of the switch 714. The switch 714 operates based on atake-over point and an input signal level as is shown in FIG. 16C. Theinput signal level is measured in the RF processing block 102. At powerup, i.e. initialization, both the RF and IF gain control signals are atthe minimum gain level and any gain adjustments that will be made arefirst applied to the RF gain control signal. As the input signal levelgets smaller, the level of the RF gain control signal increases until ithits a maximum level at the take-over point. Further increases in gainadjustment are then applied to the IF gain control signal.Alternatively, this figure shows that for small input signal levels, theRF gain is kept at a maximum level until the input signal levelapproaches the take-over point at which point the RF gain is decreasedwith an increase in the input signal level. The amount of IF gain isalso at a maximum for weak input signals, but as the input signal levelincreases, the amount of IF gain is reduced until the take-over point iscrossed at which point the IF gain level is held constant at a minimumlevel. This gain adjustment scheme allows for a maximal amount of gainto be applied earlier in the analog signal processing chain for weakinput signals and a minimal amount of gain to be applied for stronginput signals. The operating region to the right of the take-over pointcan be referred to as RF gain control mode and the region to the left ofthe take-over point can be referred to as IF gain control mode.Alternatively, the RF gain control can be disabled in embodiments inwhich the receiver uses a third party tuner that performs its own AGCregulation.

The combined loop gain and bandwidth of the lowpass filter 712determines the response time of the analog gain control block 700. Theloop gain and the bandwidth of the lowpass filter 712 can be increasedto improve AGC tracking for AM modulated television channel signals.However, increasing the loop gain and the bandwidth of the lowpassfilter 712 too much may cause the analog gain control block 700 to besusceptible to noise or to be unstable. The loop gain for the IF gainadjustment path is provided by the multiplier 716 and the amplificationfactor IF loop gain. Likewise, the loop gain for the RF gain adjustmentpath is provided by the multiplier 724 and the amplification factor RFloop gain.

When the switch 714 is configured to adjust the amount of IF gain, thefiltered gain error signal produced by the lowpass filter 712 isprovided to the IF gain adjustment path at which point it is multipliedby the IF loop gain, and then accumulated by the summer 718 and theaccumulator 720 with respect to a previous IF gain value. This can alsobe done in a negative fashion when the level of the analog IF gaincontrol signal must decrease. The accumulated IF gain value is thenprovided to the DAC 722 to produce an analog IF gain control signal. TheDAC 722 can be a 4-bit sigma-delta modulated DAC in which quantizationnoise is shifted to high frequencies in order to achieve high in-bandbit resolution with a low-resolution DAC. Accordingly, in this case ananalog low-pass filter (not shown) is also included to attenuateout-of-band noise at the output of the DAC 722, and a 4 dB back-off isused for stability purposes.

When the switch 714 is configured to adjust the amount of RF gain, thefiltered gain error signal produced by the lowpass filter 712 isprovided to the RF gain adjustment path at which point it is multipliedby the RF loop gain, and then accumulated by the summer 726 and theaccumulator 728 with respect to a previous RF gain value. This can alsobe done in a negative fashion when the level of the analog RF gaincontrol signal must decrease. The accumulated RF gain value is thenprovided to the DAC 730 to produce an analog RF gain control signal. TheDAC 730 can be a 1-bit sigma-delta modulated DAC, in which case ananalog low-pass filter (not shown) is also included to attenuateout-of-band noise at the output of the DAC 730, and a 0.45 dB back-offis used for stability purposes.

In embodiments of the receiver 100 that use a sigma-delta modulated ADCfor the ADC 106, the instability monitor 732 is employed to check thequantized output and analog QnOverRange and QnUnderRange status bits ofthe ADC 106 for detecting instability. The instability monitor 732employs a first sliding window to check the number of times that thelevel of the output of the ADC 106 hits the max and min full-scalelevels of the ADC 106. The instability monitor 732 also employs a secondsliding window to check the number of times that either the QnOverRangeor the QnUnderRange status bit is high. These status bits indicate thatthe ADC 106 may be unstable. The length of each sliding window can beset based on the ratio of the sampling rate of the ADC 106 to the centerfrequency of the desired television channel at IF as well as to providean indication of how many times the ADC 106 goes unstable during a givenperiod of the desired television channel signal. Each sliding windowalso employs a threshold with a value selected so that instability isnot detected too early, i.e. due to spurious values, or too late. Insome cases, a threshold value of 50% can be used.

When the instability monitor 732 detects instability for the ADC 106,the analog gain processing block 700 can reduce the RF or IF gainaccordingly to bring the ADC 106 to a stable state. The amount of gainadjustment is determined by the severity of the instability, which is aweighted sum of the percentage of instability indications in eachsliding window described above. When the weighted instabilityindications exceed a programmable threshold, an instability signal willbe asserted. If the RF and IF gains are reduced to the lowest levels butthe instability signal is still being asserted, then the analog gaincontrol block can output a signal to reset the ADC 106.

Although FIG. 16A shows only one analog IF gain control signal and oneanalog RF gain control signal, the analog gain control block 700 can bemodified to set the level of several variable gain amplifiers in the RFand analog processing blocks 102 and 104. The analog gain control bock700 can also be modified so that gain can be distributed amongst thedigital gain amplifier used in the video processing block 182. This canbe done using more take-over points in a similar fashion as that shownin FIG. 16B with additional take-over points being added for switchingthe amount of gain control between more than one variable gain amplifierin the IF section and for switching the amount of gain control betweenmore than one variable gain amplifier in the RF section. In other words,gain control is modified for a variable gain amplifier until it reachesa maximum/minimum setting at which point the gain control is switched toanother variable gain amplifier.

Referring now to FIG. 17A, shown therein is a block diagram of anotherexemplary embodiment of a universal television receiver 750. Theuniversal television receiver 750 employs an off-the shelf RF processingblock 752 which provides an IF multi-channel television signal 136′centered at 44 MHz (North America), 59 MHz (Japan) or 36 MHz(elsewhere). The IF multi-channel television signal 136′ includes thedesired television channel signal and at least a portion of thefrequency content of one or more adjacent television channel signals. Ingeneral, the components of the receiver 750 operate in a similar fashionas the components of the receiver 100 with differences explained below.For instance, the analog processing block 754 and the digital processingblock 756 have a similar structure and operation compared with thecorresponding blocks in the universal television receiver 100 with somechanges made to operating frequency and processing methodology.

For the analog processing block 754, switched capacitor filters are notused and so continuous time sub-sampling is not done. In addition, whilethe off-the-shelf RF processing block 752 may employ a SAW filter,coarse bandpass filters similar to filters 150 and 156 are typicallystill needed for attenuation and anti-aliasing with the requirement thatthere is sufficient rolloff to provide a sufficient amount ofattenuation (such as 72 dB for example) for overlapping signalcomponents (due to sampling) near the coarse frequency region ofinterest. However, if the RF processing block 752 provides sufficientattenuation, then no additional filtering may be necessary. Thesefilters in the analog processing block 754 have a center frequency atthe IF frequency of the RF processing block 752. The sampling rate usedfor the ADC 106 can be selected to be several times the IF frequency.For example, the sampling rate can be on the order of 288 MHz.

For the digital processing block 756, depending on the sampling rate andthe amount of downsampling that is used, some of the order of thecomponents in this block, such as a frequency rotator, and thecombination of a filter and a downsampler, may be reversed for improvedprocessing efficiency. Also, in the digital processing block 756,equalization does not have to be performed if the digital demodulator758 provides this function. In addition, analog carrier recovery isstill performed due to uncertainty in the reference frequencies used inthe frequency synthesizers in the RF and analog processing blocks 752and 754 as well as the transmitters that transmit the televisionsignals. Furthermore, the digital processing block 756 has a videoprocessing block corresponding to the video processing block 182.Accordingly, if the desired television channel signal is transmittedusing an analog broadcast standard, the digital processing block 756provides an output signal to the DAC block 110 which produces one ormore output signals 112 depending on the particular analog broadcaststandard. If the desired television channel signal is transmitted usinga digital broadcast standard, the digital processing block 756 providesa digital modulated video signal 112′ which the digital demodulator 758processes to produce a digital transport stream output 762 that can thenbe processed by an MPEG-2 decoder to produce video. This processingincludes tracking the carrier frequency of the desired televisionchannel signal. As previously mentioned, the tracking is applied to acertain frequency such as a center carrier frequency for all of thecarriers that may be used for a give digital broadcast standard.

The digital demodulator 758 can optionally provide a digital modefrequency shift feedback signal 760 to the first frequency rotator ofthe video processing block in the digital processing block 756 to adjustthe frequency shift that is applied to the processed digitized coarsechannel signal 192 so that it is centered about DC regardless of thefrequency offset error. The digital mode frequency shift feedback signal760 can be provided via software or hardware as is commonly known bythose skilled in the art to the first frequency rotator in the videoprocessing block. The digital demodulator 758 can update the value ofthe digital mode frequency shift feedback signal 760 at various timesduring operation. For instance, the digital demodulator 758 can updatethe values in the digital mode frequency shift feedback signal 760 eachtime the universal television receiver 750 is tuned to a differenttelevision channel. In other embodiments, the digital demodulator 758can also update the digital mode frequency shift feedback signal 760 toaccount for drift in the frequency offset error due to temperaturechange and the like.

Furthermore, the two-stage gain control method can be employed by theuniversal television receiver 750 in which gain control is used forvariable gain amplification in both the analog stage (i.e. analogcircuitry) and the digital stage (i.e. digital circuitry). The analoggain control techniques discussed in relation to FIGS. 15 and 16A-16Ccan be used. The digital demodulator 758 can also provide the digitalgain control signal 400, as described below.

Referring now to FIG. 17B, shown therein is a block diagram of anexemplary embodiment for the digital demodulator 758. Generally, thedigital demodulator 758 includes a demodulator block 770, an errorcorrection block 772 and a digital gain control block 774. This generalrepresentation covers any digital demodulator. For instance, for a DVB-Tdigital demodulator, the demodulator block 770 is an OFDM demodulator,and the error correction block 772 includes a Viterbi decoder and aReed-Solomon decoder. Those skilled in the art are familiar with theimplementation of the demodulator block 770, the error correction block772 and the digital gain control block 774 for a given digitaltelevision broadcast standard.

The demodulator block 770 demodulates the video information 112′provided by the video processing block of the digital processing block756. The demodulator block 770 can also lock to the carrier frequency ofthe desired television channel signal and can optionally generate thedigital mode frequency shift feedback signal 760 so that the videoprocessing block in the digital processing block 756 can compensate forfrequency offset errors when operating in digital operation mode. Theoutput of the demodulator block 770 is then processed by the errorcorrection block 772 to correct for any errors in the digital televisionchannel information and produce a digital transport stream 762. In somecases, the error correction block 772 can also produce the signal B2 tosignify that the desired television channel signal has been properlydemodulated and communicate signal B2 back to the control block 190 asexplained previously with respect to FIG. 7.

The digital gain control block 774 generates the digital gain controlsignal 400 based on the signal quality of the input data to the digitaldemodulator 758. For instance, the digital gain control block 774, inone implementation, compares the level of the input data to the digitaldemodulator 758 with a desired level, and generates an appropriate valuefor the digital gain control signal 400 so that this input signal iseither amplified or attenuated by the video processing block to achievethis level. The digital gain control block 774 is configured todetermine this amount of amplification or attenuation, as the case maybe, based on the input signal before it is processed by the demodulatorblock 770. Alternatively, the digital gain control block 774 cangenerate the digital gain control signal 400 by measuring the signalquality of the input 112′ using a digital metric and adjust the value ofthe digital gain control signal 400 to ensure that a proper signalquality is achieved. This can also include using a gain coefficienttable as was described in relation to FIG. 15.

Referring now to FIG. 18, shown therein is another embodiment of areceiver 800. The receiver 800 includes an analog processing block 802,ADC 106, a digital processing block 804, a FIFO block 806 and DAC block110. The receiver 800 receives an IF signal 808 from a third-partytelevision tuner and processes this signal to provide output signals 112or 112′ depending on whether the television broadcast standard isdigital or analog. The input to the DAC block 110 is a digitalrepresentation of a video output for an analog television broadcaststandard which is provided to the FIFO block 806 that generates theoutput signal 112′ in a digital format. A similar arrangement can beused for non-modulated audio as the SIF output is at an IntermediateFrequency (IF).

The IF signal 808 includes the desired television channel signal andother television channel signals and in conventional receivers isgenerally followed by a SAW filter and a fixed-gain amplifier tocompensate for the loss in the SAW filter. However, in the receiver 800,the SAW filter and fixed-gain amplifier are not required. Rather, thereceiver 800 takes advantage of the differences in the standards usedfor intermediate-frequencies throughout the world: 44 MHz in NorthAmerica, 59 MHz in Japan and 36 MHz for most of the remaining world.Accordingly, the components of the receiver 800 operate in asubstantially similar manner as was described for the correspondingcomponents in the receiver 750, with some changes to accommodate thethird party television tuner that provides the IF signal 808. Forinstance, the analog processing block 802 includes an optionalattenuator and a variable-gain amplifier for signal level control. Inalternative implementations, the ADC 106 is implemented as a bandpasssigma-delta ADC with its input centered at one of the aforementioned IFfrequencies. Otherwise the analog processing block 802 and the digitalprocessing block 804 operate as was described for the correspondingblocks in the receiver 750. The receiver 800 can also apply the gaincontrol techniques of FIGS. 15 and 16A-16C with modifications made, asdescribed previously, in the event that the third party tuner does notaccept an RF gain control signal. The FIFO block 806 is used to regulatethe output data flow to a downstream digital element. It should be notedthat the other receiver embodiments shown herein can employ a similarFIFO block for this purpose.

It should be noted that there can be instances in which the clock orother operational frequencies used in the receiver embodiments describedherein have significant energy at a frequency region which interfereswith the processing of the desired television channel signal thuscompromising effective SNR. This can occur during the process ofconverting analog signals to a digital representation. For example, witha sampling rate of 288 MHz and the desired television channel centeredat 36 MHz, signals at either of 252 MHz or 324 MHz would be aliased intothe desired television channel. It should be noted that the term desiredtelevision channel refers to the frequency band that includes thefrequency content of the desired television channel signal. Althoughthese frequencies are far from the input frequency, the wideband natureof television systems implies that signal power could be present there.

There can also be coupled signals that also interfere with the desiredtelevision channel signal. For example, in any combined RF/mixed-signalsystem, the problem of interfering signals within the chip is a constantchallenge. Internal oscillators, clocks, and circuitry can generate aplethora of frequencies any one of which may create interference on itsown or when combined with other frequencies within the chip. Theresulting signals, often referred to as spurs, are conventionally onlyaddressed in silicon.

Conventional television tuners have some degree of filtering whichserves to reduce power at the high aliasing frequencies. However, thisfiltering has no effect on local-oscillator leakage from the tuner orthe other spurious signals just described. Although the leakage powercan be as large as some television channels, it is conventionallyreduced by a combination of SAW filter(s) and anti-aliasing filtersbefore the ADC 106. However, no SAW filters are used and a minimum ofanti-alias filtering is employed in the various embodiments of theanalog processing block described herein; thus, the local-oscillatorsignal is present at full strength and may be at an aliasing frequency(i.e. a frequency that is aliased onto the frequency range that containsthe desired television channel signal during processing, i.e. aliasedonto the desired television channel signal). In this regard, even asingle tone may be problematic for television signals transmittedaccording to certain broadcast standards.

To mitigate the effects of these different types of interference, it canbe assumed that interference occurs at certain frequencies and if thesefrequencies coincide with the desired television channel, then thecontrol block 190 can shift clock and sampling rates so that theinterference is no longer in the frequency region where the desiredtelevision channel signal is being processed and no longer adverselyaffects the processing of the desired television channel signal. Theadjustment in sampling rate frequency shifts an aliased version of theinterferer away from the desired television channel. Resampling ratiosare used to compensate for the adjusted sampling rate as describedbelow. The shift in clock frequency affects the frequency value of themixing signals and the sampling rate. However, because of the use ofcoarse filtering, coarse channel signals, and the carrier frequencyrecovery performed in the video and audio processing blocks, the ADC,digital processing block, DAC and the LO of the various embodimentsdescribed herein can operate together to allow the sampling rate of theentire receiver to vary. Normally, in conventional receivers, thesampling rate is fixed. However, the technique of variation in samplingrate can be employed by the receivers described herein to allow for theavoidance of potential alias signals and spurious signals generated bythe various clocking domains by shifting the frequency regions that foldback or alias, due to the nature of sampling, onto the desiredtelevision channel. Other shifts can also be employed to othercomponents of the receiver to further deal with interferers and this isdiscussed further below.

Referring now to FIG. 19A, shown therein is an alternative embodiment ofa universal receiver 900 that employs shifts in clock frequency andsampling rate to avoid interference that can be aliased onto the desiredtelevision channel. Although this technique is described with referenceto the receiver 900, the technique can also be used with the otherreceiver embodiments described herein. The receiver 900 comprises ananalog processing block 902, an ADC 106, a digital processing block 904,a DAC block 110 and a FIFO 906. These blocks generally operate in asimilar fashion as the corresponding blocks in receivers 750 and 800with the additional feature of aliasing avoidance sampling rateadjustment described below. The receiver 900 also includes a variablePhase Lock Loop (PLL) 916, and the control block 190.

To avoid or otherwise mitigate the effects of aliased interference, thesampling rate of the receiver 900 can be changed during operation. Sincethe sampling rate of the ADC 106 is not fixed, the digital processingblock 904 is configured to operate at a sampling rate that correspondsto the modified sampling rate of the ADC 106. This is also true for theDAC block 110. The entire digital signal path maintains a consistentclock rate. As noted earlier, in the description of the receiver 100,near the input and output of the digital processing block 904 there arepolyphase filters. In particular, the video processing block 182includes polyphase filters 352 and 368, the audio filtering blocks 184and 186 include polyphase filters 554 and 566, and the audio processingblock 188 includes polyphase filters 606, 614, and 628. This is alsotrue for the digital processing block 904. These polyphase filters 352,368, 554, 566, 606, 614 and 628 are used to change the effectivesampling rate that is employed so that signal sampling rates internallymaintain the proper ratio with respect to physical clock rates to ensurethat the filtering and processing characteristics within these blocksare consistent regardless of the actual physical sampling rates that areused. Accordingly, the polyphase filters 352 and 554 at the inputs ofthe video processing block 182, the audio filtering blocks 184 and 186and the audio processing block 188 provide a first conversion in theeffective sampling rate so that processing within these blocks occurs asif the physical sampling rate was never changed. The polyphase filters368, 566, 606, 614 and 628 at the outputs of these blocks 182-188 thenapply a second conversion in the effective sampling rate to convert backto the physical sampling rate that is used by the DAC block 110 in orderto properly generate the analog output signals. Alternatively, adifferent output rate can be used if a standard digital output data ratehad to be accommodated. Accordingly, this technique of interferenceavoidance includes using a first input resampling ratio for the “input”polyphase filters 352 and 554 to transform the adjusted sampling rate toa nominal processing rate that was otherwise going to be used for theprocessing elements between the input and output polyphase filters andusing at least one output resampling ratio for the “output” polyphasefilters 368, 566, 606, 614 and 628 to transform the nominal processingrate to the adjusted sampling rate or another sampling rate (moreoutputs resampling ratios can be used if the different outputs of thereceiver 900 are to be provided at different rates).

Accordingly, the digital processing block 904 is configured to applyresampling ratios to compensate for the adjusted sampling rate that isused to avoid aliased interferers. This technique can be applied toaliased versions of other interferers and is not restricted tointerferers generated by the local oscillator of the tuner that is usedin conjunction with the receiver. For example, harmonics of the IFpicture carrier of the desired television channel signal as well asdistortion products of various components of the receiver arepredictable sources of interferers whose frequency, and the frequency ofthe aliased versions thereof are known a priori and can therefore behandled with this technique.

In addition, it should be noted that this aliasing avoidance techniquecan be used with any tuner implementation and not just the tunerimplementations that are described herein. For example, this techniqueis applicable to single conversion tuners, dual-conversion tuners andsuper-heterodyne tuners.

The technique of shifting the sampling rate to avoid aliasedinterference involves several aspects. Firstly, because the physicalsampling rate has changed, but the desired television channel signalremains at the same input frequency relative to the ADC 106, aftersampling, the desired television channel signal is now further offsetfrom its expected location in terms of normalized frequency. As has beendescribed previously, the architecture employed for the various receiverembodiments described herein allows for some shift in the frequencylocation of the desired television channel. The frequency change isdetermined by the control block 190. The second aspect involves alteringthe effective sampling rate that is employed by the polyphase filters352 and 554 in order to frequency translate/transform signals forprocessing by subsequent fixed-width filtering stages when thistechnique is used with the frequency resampling technique discussedearlier for handling different channel bandwidths for differentbroadcast standards. To accomplish this, a first or input resamplingratio is applied, which is the ratio of the original sampling rate tothe new desired sampling rate, to reflect the altered relative valuesbecause of the new physical sampling rate. Thirdly, at the output of thedigital processing block 182, the sampling rate is returned to itsoriginal state prior to signal conversion to the analog domain by theDAC block 110 by applying a second or output resampling ratio. Thesecond ratio is typically the inverse of the first ratio, however,another value can be used for the second resampling rate if the outputrate is desired to be at another different rate.

Table 1 illustrates how the resampling rates are modified to provideconsistent operation when the physical sample rates of the ADC and DACare changed in order to provide various modes of interference avoidancewhile configured to receive an NTSC television signal, for example. Allvalues in Table 1 are in MHz except for the ratios F_(s3)/F_(s2) andF_(s5)/FS_(s4). It can be seen that as the sample rate of the ADC 106changes to avoid aliasing of interferers into the desired televisionchannel, the input to the polyphase filter 352, after being downsampledby a factor of 8, also changes. However, the sample rate for the outputof the video polyphase filter 352 is held constant because the amount ofresampling provided by the video polyphase filter 352 is changed in asimilar manner as the change in the sample rate of the ADC 106. Thisallows the components between the video polyphase filter 352 and thevideo polyphase filter 368 to operate as if there had been no change tothe sample rate of the ADC 106. The video polyphase filter 368 thenapplies a corresponding inverted resampling ratio (the correspondingnumbers in rows 4 and 6 are the inverse of one another) so that thesample rate of the output of the video polyphase filter 368 matches thesample rate of the input of the video polyphase filter 352. Likewise,the sample rate of the input to the DAC 372 is the same as the samplerate of the output of the ADC 106. However, as mentioned, in some casesthe video polyphase filter 368 applies a different resampling ratio sothat the sampling rate of the input to the DAC 372 is different from thesample rate of the output of the ADC 106. This change in sample rate toavoid aliasing can be done independently or in addition to the change insample rate for multi-channel processing described with respect to FIG.10 depending on the particular receiver architecture in which thetechnique of sampling rate adjustment is employed.

TABLE 1 Exemplary Values for Resampling Ratios (all values in MHz) ModeSample Rate 1 2 3 4 5 Output of ADC 106 (F_(s1)) 280 284 288 292 296Input to video polyphase filter 35.0 35.5 36 36.5 37.0 352 (F_(s2))Output of video polyphase 15.16 15.16 15.16 15.16 15.16 filter 352(F_(s3)) F_(s3)/F_(s2) 0.4331 0.4270 0.4211 0.4153 0.4097 Input to videopolyphase filter 15.16 15.16 15.16 15.16 15.16 368 (F_(s4)) Output ofvideo polyphase 35.0 35.5 36.0 36.5 37.0 filter 368 (F_(s5))F_(s5)/F_(s4) 2.3087 2.3417 2.3747 2.4077 2.4406 Input to DAC 372(F_(s6)) 280 284 288 292 296

In an exemplary embodiment, in order to shift the physical clock andsampling rates, the receiver 900 employs the variable PLL 916, which cangenerate several fixed output frequencies based on a sampling ratecontrol signal received from the control block 190. To determine whethera sampling rate offset is required, the control block 190 considers thefrequencies used by all of the local oscillators that are employed bythe receiver 900 as well as the frequencies used by the localoscillators that are employed in circuits which are connected to thereceiver 900, such as an RF front end tuner. The control block 190considers these local frequencies and harmonics of these localoscillator frequencies to be interferers. The control block 190 thenconsiders the aliasing bands associated with a plurality of samplingrates that can be selected from. The aliasing bands are determined basedon the desired television channel that is being digitized and thesealiasing bands basically occur at integer multiples of the sampling rateplus or minus the IF frequency range that corresponds to the desiredtelevision channel (see FIGS. 19C and 19D for an example of the aliasingbands at the first integer multiple). When the control block 190determines that at least one interferer resides in an aliasing band whenusing the nominal sampling rate, the control block 190 checks for othersampling rates for which the aliasing bands do not include an interfererand then calculates the required values for the resampling ratios.Accordingly, while the amplitude of the interferer is not known apriori, it is assumed that the presence of an aliased interferer withinthe desired television channel will have a detrimental effect on thequality of the desired television channel signal and should be removed.

For example, the variable PLL 916 can generate a clock signal at fivefixed output frequencies: 280 MHz, 284 MHz, 288 MHz, 292 MHz and 296MHz. The nominal operating frequency can be set to the middle of thefive values, i.e. 288 MHz. When the control block 190 determines that aninterferer is in an aliasing band of the 288 MHz sampling rate, thecontrol block 190 can instruct the variable PLL 916 to generate amodified clock signal for which the interferer is not in an aliasingband. The modified clock signal is used to provide the adjusted samplingrate. The modified clock signal is then provided to the ADC 106, thedigital processing block 904 and the DAC block 110.

In an exemplary embodiment shown in FIG. 19B, the variable PLL 916includes a Phase/Frequency Detector (PFD) 920, a charge (Q) pump 922, aVoltage Controlled Oscillator (VCO) 924 and a division control block926. The PFD 920 receives a reference clock signal and a frequencydivided version of an oscillation signal that is output by the VCO 924.The PFD 920 determines the error in phase for the frequency dividedsignal with respect to the reference signal, and provides this phaseerror to the charge pump 922. The charge pump 922 then generates anoutput voltage to track the desired output frequency of the variable PLL916. The VCO 924 receives the output voltage of the charge pump 922 andgenerates the oscillation signal. The frequency of the oscillationsignal is then shifted to a lower rate by the division control block 926according to the sampling rate control signal provided from the controlblock 190 based on a desired sampling rate. The division control block926 then provides the desired clock signal for sampling at the desiredsampling rate.

Referring now to FIG. 19C, shown therein is a spectral plot showing theeffects of aliased interference when the aliasing avoidance technique isnot employed. In this example, a nominal sampling rate of 288 MHz isemployed for analog-to-digital conversion and the desired televisionchannel signal is a European 8 MHz wide television channel centered at36 MHz. The ADC 106 is also centered at 36 MHz which is ⅛^(th) of thesampling rate. An interfering signal is present at 323 MHz, which couldbe a Local-Oscillator (LO) signal from a third party tuner (i.e. RFprocessing block) for example, and the aliased version of this signalappears 1 MHz away from the center of the coarse region of interest atIF (as the interferer is 1 MHz away from the center of the aliasingband), which reduces the effective SNR of the desired television channelsignal. However, by changing the frequency of the clock signal providedby the variable PLL 916, the sampling rate can be altered as shown inFIG. 19D. For instance, if the sampling rate is increased to 296 MHz,the center of the digitized coarse region of interest is now at 37 MHz,and the desired television channel remains between 32 and 40 MHz.However, the aliased version of the interfering signal at 323 MHz is now10 MHz away from the center of the digitized coarse region of interestand thus falls at an equivalent frequency of 27 MHz. Alternatively, thesampling rate can be changed to 292 MHz in which case the digitizedcoarse region of interest is now centered at 36.5 MHz. The desiredtelevision channel remains between 32-40 MHz and but the aliasedinterferer is moved to a frequency of 31 MHz. Alternatively, thesampling rate can be changed to 280 MHz which shifts the aliasedinterferer to a frequency of 43 MHz.

In an alternative, in addition to changing the sampling rate, anothermodification that can be made to move an aliased interferer away fromthe desired television channel is to also shift the local oscillatorfrequency. For instance, even if the sampling rate is changed, analiased interferer may lie on the edge of the desired televisionchannel. In this case, since the number of sampling rates may belimited, in order to provide an additional frequency shift to move thealiased interferer away from the desired television channel, thefrequency of the local oscillator can be slightly moved when the localoscillator is the cause of the interferer. The effect of shifting thefrequency of the local oscillator in this way is also a shift in the IFfrequency of the desired television channel. However, this shift can betolerated due to the coarse nature of the filtering that is employed bythe receiver 900 as well as the frequency locking that is employed bythe video and audio processing blocks, which were described for thereceiver 100.

In another alternative, in addition to changing the sampling rate,another modification that can be made to move an aliased interferer awayfrom the coarse frequency region of interest is to calculate the offsetof the center frequency of the desired television channel within thecoarse frequency region (i.e. the capture bandwidth of the ADC 106).This offset is available based on the frequency tracking that is done bythe video processing block 182 as described with reference to FIG. 10.The sampling rate can then be adjusted accordingly so that the aliasedinterferer does not overlap with the desired television channel eventhough the aliased interferer may overlap with a portion of the coarsefrequency region of interest.

In another embodiment, the receiver can be modified to addressinterferers that are not aliased such as those interferers due todistortion, coupling and the like with respect to the Local Oscillator(LO) used in the mixing stage. Accordingly, these types of interferersinclude any distortion components that can end up at the desiredtelevision channel, at the image from the LO frequency or at the IFfrequency. These can be due to harmonic, intermodulation, or mixingresults and can be any combination of the three. For instance,interference may be due to a distortion product in which the localoscillator of the RF processing block (i.e. tuner) is one of thefrequency sources. Alternatively, the interference may result when anycombination of the video carrier and/or audio carrier of the desiredtelevision channel signal has generated a strong intermodulation tone orharmonic product which combine at the image frequency of the LO and arethen placed onto the desired television channel signal.

The receiver can be modified by shifting the LO frequency of the mixingstage with or without sampling rate adjustment. The LO frequency canfirst be shifted, without employing the technique of adjusting thesampling rate described above, to frequency shift interferers away fromthe desired television channel. Due to this frequency shift, the desiredtelevision channel may move, but this poses no difficulties forfrequency shifts up to a certain amount due to the use of coarsefiltering that is employed in the RF and analog processing blocks.Receiver architectures with traditional SAW filters cannot accommodateany such shift in LO frequency since SAW filters are very frequencyspecific and necessitate the precise placement of the desired televisionchannel signal to ensure that it is properly filtered without losing anyinformation. However, the various embodiments of the receiverarchitectures described herein use imprecise or coarse filters that havecoarse pass bands that are wider than the bandwidth of the desiredtelevision channel and can therefore accommodate a shift in frequency ofthe desired television channel at IF at the output of the tuner (i.e.output of the RF processing block) and at the input to the demodulator(i.e. input to the analog processing block).

With this technique, the local oscillator provides a variableoscillation frequency that can be shifted depending on the location ofthe interferer. As the LO frequency is shifted, the location of theinterferer tone will be moved. If the difference in frequency betweenthe interferer and the location of the desired television channel issmall then merely adjusting the LO frequency will result in a sufficientchange to move the interferer out of the desired television channelwhile keeping the desired television channel within the confines of thecoarse pass bands of the filters used in the analog processing block.The amount of frequency shift can be known a priori based on the type ofinterferer. For instance, if the interferer is due to a distortionproduct of at least one of the picture carrier and/or audio carrier ofthe desired television channel signal combined with the frequencylocation of the local oscillator or its image, then the amount of theshift in the variable oscillation frequency is that which is required toshift the interferer away from the desired television channel or itsimage.

For example, referring now to FIG. 20A shown therein is a spectral plotillustrating interference of a desired television channel due todistortion. In this case prior to mixing by the local oscillator, thepicture and audio carriers of the desired television channel signal havefrequencies of 76.25 MHz and 81.75 MHz respectively. The localoscillator frequency is 115.15 MHz and after mixing the picture andaudio carriers have frequencies of 38.9 and 33.4 MHz respectively (i.e.LO frequency-picture carrier frequency and LO frequency-audio carrierfrequency). However, after mixing a first interferer is also present at37.35 MHz. This interferer is due to the mixing of distortion, i.e. aharmonic of the picture carrier at 152.5 MHz (i.e. 2*76.25 MHz) hasmixed with the LO frequency. There is also a second interferer presentat 42.85 MHz which is due to the intermodulation of the picture andaudio carriers mixing with the LO. Note that that other distortion (notshown) due to a harmonic of the audio carrier (i.e. 2*audio carrierfrequency) or other combinations of the picture and audio carrierfrequencies (i.e. audio carrier frequency-picture carrier frequency) mayget mixed into the desired television channel by the LO depending on thefrequency value of the distortion and the LO frequency. In this case thefirst interferer at 37.35 MHz may be problematic. It should also benoted that in this case for the ADC, the sampling rate is 288 MHz, thesampling band center is at 36 MHz, the lower limit of the sampling bandis 31 MHz and the upper band of the sampling limit is 41 MHz. This isshown as the ADC sampling band in FIG. 20A.

Referring now to FIG. 20B, shown therein is a spectral plot illustratingthe avoidance of the distortion interference of FIG. 20A by applying alocal oscillator frequency shift. In this case, the picture carrier,audio carrier and distortion are at the same frequencies of 76.25 MHz,81.75 MHz and 152.5 MHz prior to mixing, however the LO frequency hasnow been shifted slightly to 113.65 MHz. Accordingly, after mixing thepicture and audio carriers have frequencies of 37.4 and 31.9 MHzrespectively while the first interferer now has a frequency of 38.85 MHzand the second interferer now has a frequency of 44.35 MHz. Accordingly,in this case both of the interferers have now been frequency shiftedaway from the desired television channel by applying a small frequencyshift to the LO frequency such that no interferer directly lies withinthe desired television band. Also, the sampling properties of the ADChave not been changed.

In those cases in which a much larger shift is needed in the variableoscillation frequency of the LO such that the desired television channelis no longer within the coarse pass bands of the analog processingblock, the technique of adjusting the sampling rate is also used toensure that the interferer is shifted away from the desired televisionchannel while still capturing the desired television channel within thecoarse pass bands of the coarse filters employed by the analogprocessing block. The sampling rate adjustment moves the center of theband of interest (i.e. that which is captured by the ADC) 106) andtherefore accommodates for the resulting frequency shift in the desiredtelevision channel due to changing the oscillator frequency.Accordingly, when the LO frequency has been changed by such an amountthat the desired television channel moves out of the capture bandwidthof the ADC 106, the adjustment in sampling rate can be used to move thecenter of the capture bandwidth of the ADC 106 by an appropriate amountso that it now includes the desired television channel and excludes theinterferer.

For example, referring now to FIG. 21A shown therein is a spectral plotillustrating interference of a desired television channel due todistortion. In this case prior to mixing by the local oscillator, thepicture and audio carriers of the desired television channel signal havefrequencies of 69.25 MHz and 74.75 MHz respectively. The localoscillator frequency is 108.2 MHz and after mixing the picture and audiocarriers have frequencies of 38.9 and 33.4 MHz respectively (i.e. LOfrequency-picture carrier frequency and LO frequency-audio carrierfrequency). However, after mixing interferer 3 is present at 30.35 MHz.This interferer is due to the mixing of distortion, i.e. a harmonic ofthe picture carrier at 138.5 MHz (i.e. 2*69.25 MHz) has mixed with theLO frequency. There is also interferer 4 present at 35.85 MHz which isdue to intermodulation of the picture and audio carriers mixing with theLO. In this case interferer 4 at 35.85 MHz may be problematic. It shouldalso be noted that in this case for the ADC, the sampling rate is 288MHz, the sampling band center is at 36 MHz, the lower limit of thesampling band is 31 MHz and the upper limit of the sampling band is 41MHz.

Referring now to FIG. 21B, shown therein is a spectral plot illustratingthe avoidance of the distortion interference of FIG. 21A by applying alocal oscillator frequency shift. In this case, the picture carrier,audio carrier and distortion are at the same frequencies of 69.25 MHz,74.75 MHz, 138.5 and 144 MHz prior to mixing, however the LO frequencyhas now been shifted slightly to 110.3 MHz. Accordingly, after mixingthe picture and audio carriers have frequencies of 41 and 35.5 MHzrespectively while interferer 3 now has a frequency of 28.25 MHz andinterferer 4 has a frequency of 33.75 MHz. Accordingly, in this caseboth interferers have also been frequency shifted away from the desiredtelevision channel by applying a small frequency shift to the LOfrequency such that no interferer directly lies within the desiredtelevision band. However, the television band has also shifted such thatit doesn't correspond with the sampling band of the ADC. Accordingly, inthis case, the sampling properties of the ADC are changed such that thesampling rate is now 296 MHz, the sampling band center is now at 37 MHz,the lower limit of the sampling band is now at 32 MHz and the upperlimit of the sampling band is now at 42 MHz. This allows the shifteddesired television channel to be properly digitized without loss ofinformation. It should be noted that the amplitudes and frequencyspacing shown in FIGS. 20A, 20B, 21A and 21B are not shown to scale.

It should be noted that various aspects of processing methodology andcorresponding structure have been provided herein for several differentembodiments of a television receiver. Processing techniques andcorresponding structure have been described for processing widebandtelevision channel signals to obtain the video and audio information ofa desired television channel signal that can be transmitted according toa variety of analog or digital broadcast standards. This includesapplying resampling that is configurable based on the particularbroadcast standard so that a main fixed video or main fixed audio filtercan be used to process television channel signals of various bandwidths.Processing techniques and corresponding structure have also beendescribed for employing variable gain control that includes acombination of analog and digital variable gain control. Severaldifferent techniques for determining how the gain is varied duringoperation are provided herein. Processing techniques and correspondingstructure have also been described for employing phase noise reductionto compensate for any phase noise in the desired television channelsignal when transmitted under an analog broadcast standard. Processingtechniques and corresponding structure have also been described forusing various “coarse techniques” for filtering or mixing as well asusing frequency tracking to accommodate various changes due totransmission frequency or hardware as described herein. Processingtechniques and corresponding structure have also been described forinterference avoidance based on adjusting sampling rate, shiftingcertain oscillation frequencies or both adjusting sampling rate andshifting certain oscillation frequencies. Processing techniques andcorresponding structure have also been described for compensating forovermodulation. Various embodiments for each of these processingtechniques and corresponding structure have been described herein. Itshould be noted that these processing techniques and correspondingstructure can all be used together in one embodiment, or varioussub-combinations of these processing techniques and correspondingstructure can be used as described herein when appropriate, or one ormore of these techniques can be used in other receiver architectureswhen appropriate (i.e. “when appropriate” means that the end result is aworking embodiment).

It should be noted that the filtering and downsampling that is performedby various blocks in the television receiver 100 may be realized bycascading several filters and downsamplers in series. This results inimproved realization efficiency and greater processing efficiency sincefilters with a smaller number of filter taps can be used. Further, itshould be noted that the sampling rates, the degree of downsampling andthe sequence order of the various filters, downsamplers and frequencyrotators can be adjusted for more efficient implementation. Also, itshould be understood that the digital processing block 108 isimplemented as a combination of an application specific integratedcircuit along with a digital signal processor, with registers and memoryand the like. Accordingly, the functionality of the blocks in thedigital processing block 108 is implemented using a combination ofhardware and software. It should further be understood that thesevarious blocks in the digital processing section can be implemented witha different structure, either in hardware or software, from that shownherein as long as the same functionality is maintained. Similarly,modifications can be made to the RF and analog processing blocks as longas the basic functionality is maintained.

It should also be noted that the various embodiments of the receiversdescribed herein are generally configured to process analog televisionbroadcast standards comprising NTSC, SECAM, and PAL and digitaltelevision broadcast standards comprising ATSC, DVB-T, DMB-T and ISDB-T.Also, it should be noted that the term amplification circuitry can beinterpreted to include variable gain amplifiers or amplifiers that donot have variable gain.

In one aspect, at least one of the embodiments described herein providesa television receiver for processing received television signals toprovide video and audio information for a desired television channelsignal. The television receiver comprises an analog processing block forproviding coarse filtering and amplification to a multi-channeltelevision signal to produce a first signal, the coarse filtering beingconfigured to use pass bands that are wide enough to accommodatefrequency shifts in the desired television channel signal and analogcircuitry variability; an analog to digital converter coupled to theanalog processing stage for digitizing the first signal to produce asecond signal; and a digital processing block coupled to the analog todigital converter for processing the second signal to obtain the videoand audio information for the desired television channel signal. Thereceiver is configured to track a carrier frequency of the desiredtelevision channel signal and generate and apply a frequency shiftfeedback signal to compensate for frequency shifts in the carrierfrequency.

The digital processing block can be configured to additionally apply aknown frequency shift for known frequency offset errors in the carrierfrequency.

The receiver can further comprise an RF processing block coupled to theanalog processing block for receiving broadcast television signals andproviding amplification, filtering and mixing to generate themulti-channel television signal, wherein the RF processing blockcomprises a mixing stage having an oscillator with a coarse step size ora fine step size, and the digital processing block is further configuredto compensate for the step size of the oscillator when generating theknown frequency shift.

The analog processing block comprises at least one coarse bandpassfilter or at least one coarse low pass filter for providing analogfiltering.

If the television receiver comprises an RF processing block, the mixingstage can comprise elements for providing an additional level of analogfiltering.

The digital processing block can also be further configured to provideequalization for compensating for non-ideal analog filtering in at leastone of the analog and RF processing blocks.

The first signal is a coarse channel signal, the second signal is adigitized coarse channel signal, and the digital processing block isfurther configured to process the digitized coarse channel signal toprovide a processed digitized coarse channel signal, and the digitalprocessing block comprises a video processing block having a frequencytracking loop for tracking the carrier frequency of the desiredtelevision channel signal. The frequency tracking loop comprises a firstfrequency rotator for shifting the frequency content of the processeddigitized coarse channel signal to the baseband based on the knownfrequency shift and additionally the frequency shift feedback signal foranalog broadcast transmission standards; a video filter stage forfiltering the output of the first frequency rotator; and a digitalvariable gain amplifier for amplifying the output of the video filterstage.

The receiver can further comprise a digital demodulator coupled to thedigital processing block for processing the output of the digitalvariable gain amplifier and tracking the carrier frequency of thedesired television channel signal to provide a digital transport streamin accordance with a digital television broadcast standard.

In this case, the digital demodulator can further be configured togenerate and provide the frequency shift feedback signal to the firstfrequency rotator when the desired television channel signal isbroadcast according to a digital television broadcast standard.

The frequency tracking loop further comprises a second frequency rotatorcoupled to the digital variable gain amplifier for frequency shiftingthe output of the digital variable gain amplifier; and a picture carrierrecovery block coupled to the first and second frequency rotators, thepicture carrier recovery block being configured to receive the output ofthe second frequency rotator, generate and provide the frequency shiftfeedback signal to the first frequency rotator and provide the videoinformation of the desired television channel signal for analogbroadcast transmission standards.

The picture carrier recovery block comprises a carrier recovery filterconfigured to filter the output of the second frequency rotator toproduce a filtered picture carrier signal; a first phase rotatorconfigured to apply phase adjustment to the filtered picture carriersignal to produce a phase-adjusted filtered picture carrier signal; anda carrier recovery block coupled to the first phase rotator and thefirst frequency rotator, the carrier recovery block being configured toprocess the phase-adjusted filtered picture carrier signal to compensatefor phase noise and produce a phase control signal that is provided tothe first phase rotator to control the amount of the phase adjustment,the carrier recovery block further being configured to generate thefrequency shift feedback signal.

The carrier recovery block includes a phase-frequency detectorconfigured to receive an input phase signal based on the output of thesecond frequency rotator, generate a phase error signal by comparing theinput phase signal with a reference phase signal, and generate afrequency error signal based on the phase error signal; a summerconfigured to sum the frequency error signal and a version of the phaseerror signal to produce an adjusted frequency error signal; a frequencyloop amplifier configured to provide an amplified frequency error signalbased on the adjusted frequency error signal; and a frequency oscillatorblock configured to generate the frequency shift feedback signal basedon the amplified frequency error signal.

The frequency oscillator block comprises a frequency accumulatorconfigured to update a current frequency based on the amplifiedfrequency error signal to produce a frequency adjusted signal; and afrequency clipping block configured to specify upper and lower limitsfor the frequency adjusted signal, and when the frequency adjustedsignal exceeds one of the limits, the frequency clipping block isconfigured to limit the frequency adjusted signal to that limit. Thefrequency shift feedback signal is generated based on the output of thefrequency clipping block.

The carrier recovery block further comprises a low pass filterconfigured to filter the phase error signal to produce a filtered phaseerror signal; and a phase loop amplifier configured to amplify thefiltered phase error signal to produce the version of the phase errorsignal.

The digital processing block further comprises an audio filtering blockhaving a second frequency tracking loop configured to extract an audiocarrier frequency of the desired television channel signal for analogtelevision broadcast standards and the audio filtering block isconfigured to apply a second known frequency shift to compensate for aknown frequency offset in the audio carrier frequency.

The second frequency tracking loop comprises a third frequency rotatorfor shifting the frequency content of the processed digitized coarsechannel signal to the baseband; an audio filter stage for filtering theoutput of the third frequency rotator; a frequency demodulator coupledto the audio filter stage for demodulating the output of the audiofilter stage and producing a first intermediate audio signal; and anaudio IF carrier recovery block that is configured to track the audiocarrier signal that corresponds to the desired television channel signalbased on one of the first intermediate audio signal and a sound IFcarrier recovery signal, and to produce an audio frequency shiftfeedback signal which is provided to the third frequency rotator.

In an alternative, the digital processing block further comprises anaudio filtering block that is configured to process the output of thefirst frequency rotator to extract an audio carrier frequency of thedesired television channel signal for analog television broadcaststandards, wherein the processing is based on the phase control signal.

In this alternative, the audio filtering block comprises a thirdfrequency rotator for shifting the frequency content of the frequencyshift feedback signal to the baseband; an audio filter stage forfiltering the output of the third frequency rotator; a frequencydemodulator for demodulating the output of the audio filter stage andproducing a first intermediate audio signal; and an audio IF carrierrecovery block that is configured to track the audio carrier signal thatcorresponds to the desired television channel signal.

In another aspect, at least one of the embodiments described hereinprovides a method of processing received television signals in atelevision receiver to provide video and audio information for a desiredtelevision channel signal. The method comprises providing coarsefiltering and amplification to a multi-channel television signal toproduce a first signal, the coarse filtering being configured to usepass bands that are wide enough to accommodate frequency shifts in thedesired television channel signal and analog circuitry variability inthe television receiver; digitizing the first signal to produce a secondsignal; and processing the second signal to obtain the video and audioinformation for the desired television channel signal by tracking acarrier frequency of the desired television channel signal andgenerating and applying a frequency shift feedback signal to compensatefor frequency shifts in the carrier frequency.

The method further comprises additionally applying a known frequencyshift for known frequency offset errors in the carrier frequency.

The method can comprise utilizing an RF processing block for receivingbroadcast television signals and providing amplification, filtering andmixing to generate the multi-channel television signal, the RFprocessing block comprising a mixing stage having an oscillator with acoarse step size or a fine step size, and the method further comprisescompensating for the step size of the oscillator when generating theknown frequency shift.

The step of providing coarse filtering and amplification comprisesutilizing at least one coarse bandpass filter or at least one coarse lowpass filter for providing analog filtering.

When using an RF processing block, the method further comprisesproviding the mixing stage with elements for providing an additionallevel of analog filtering.

The processing step can comprise providing equalization for compensatingfor non-ideal analog filtering.

The first signal is a coarse channel signal, the second signal is adigitized coarse channel signal, and the method further comprisesprocessing the digitized coarse channel signal to provide a processeddigitized coarse channel signal; shifting the frequency content of theprocessed digitized coarse channel signal to the baseband based on theknown frequency shift and additionally the frequency shift feedbacksignal for analog broadcast transmission standards; filtering theshifted frequency content of the processed digitized coarse channelsignal with a video filter stage to produce a filtered signal; andamplifying the filtered signal to produce an amplified signal.

The method can further comprise using a digital demodulator to processthe amplified signal and track the carrier frequency of the desiredtelevision channel signal to provide a digital transport stream inaccordance with a digital television broadcast standard.

In this case, the method can further comprise using the digitaldemodulator to generate and provide the frequency shift feedback signalwhen the desired television channel signal is broadcast according to adigital television broadcast standard.

For analog broadcast transmission standards, the method furthercomprises frequency shifting the amplified signal to produce a frequencyshifted amplified signal; generating the frequency shift feedback signalbased on the frequency shifted amplified signal and providing thefrequency shift feedback signal to the first frequency rotator; andfiltering the frequency shifted amplified signal to provide the videoinformation of the desired television channel signal.

The method can further comprise filtering the frequency shiftedamplified signal to produce a filtered picture carrier signal; applyingphase adjustment to the filtered picture carrier signal to produce aphase-adjusted filtered picture carrier signal; and processing thephase-adjusted filtered picture carrier signal to compensate for phasenoise, to produce a phase control signal that is used to control theamount of the phase adjustment, and to generate the frequency shiftfeedback signal.

The method further comprises receiving an input phase signal based onthe frequency shifted amplified signal; generating a phase error signalby comparing the input phase signal with a reference phase signal;generating a frequency error signal based on the phase error signal;summing the frequency error signal and a version of the phase errorsignal to produce an adjusted frequency error signal; amplifying thephase adjusted frequency error signal to produce an amplified frequencyerror signal; and generating the frequency shift feedback signal basedon the amplified frequency error signal.

Generating the frequency shift feedback signal comprises updating acurrent frequency based on the amplified frequency error signal toproduce a frequency adjusted signal; limiting the frequency adjustedsignal to an upper or lower limit when the frequency adjusted signalexceeds the upper or lower limit respectively; and generating thefrequency shift feedback signal based on the limited frequency adjustedsignal.

The method further comprises filtering the phase error signal to producea filtered phase error signal; and amplifying the filtered phase errorsignal to produce the version of the phase error signal.

The method further comprises using a second frequency tracking loop toextract an audio carrier frequency of the desired television channelsignal for analog television broadcast standards by applying a secondknown frequency shift to compensate for a known frequency offset in theaudio carrier frequency.

The method further comprises frequency shifting the frequency content ofthe processed digitized coarse channel signal to the baseband; filteringthe frequency shifted signal using an audio filtering stage to produce afiltered signal; demodulating the filtered signal to produce a firstintermediate audio signal; and tracking the audio carrier signal thatcorresponds to the desired television channel signal based on one of thefirst intermediate audio signal and a sound IF carrier recovery signaland producing an audio frequency shift feedback signal used in thefrequency shifting step.

In an alternative, the method further comprises processing the shiftedfrequency content of the processed digitized coarse channel signal toextract an audio carrier frequency of the desired television channelsignal for analog television broadcast standards, wherein the processingis based on the phase control signal.

In this alternative, the method further comprises frequency shifting theshifted frequency content of the processed digitized coarse channelsignal to the baseband; filtering the frequency shifted signal using anaudio filtering stage to produce a filtered signal; demodulating thefiltered signal to produce a first intermediate audio signal; andtracking the audio carrier signal that corresponds to the desiredtelevision channel signal based on the first intermediate audio signal.

It should be understood that various modifications can be made to theembodiments described herein, without departing from these embodiments,the scope of which is defined in the appended claims.

1. A television receiver for processing received television signals toprovide video and audio information for a desired television channelsignal, wherein the television receiver comprises: an analog processingblock for providing coarse filtering and amplification to amulti-channel television signal to produce a first signal, the coarsefiltering being configured to use pass bands that are wide enough toaccommodate frequency shifts in the desired television channel signaland analog circuitry variability; an analog to digital converter coupledto the analog processing stage for digitizing the first signal toproduce a second signal; and a digital processing block coupled to theanalog to digital converter for processing the second signal to obtainthe video and audio information for the desired television channelsignal, wherein the receiver is configured to track a carrier frequencyof the desired television channel signal and generate and apply afrequency shift feedback signal to compensate for frequency shifts inthe carrier frequency.
 2. The television receiver of claim 1, whereinthe digital processing block is configured to additionally apply a knownfrequency shift for known frequency offset errors in the carrierfrequency.
 3. The receiver of claim 2, wherein the receiver furthercomprises an RF processing block coupled to the analog processing blockfor receiving broadcast television signals and providing amplification,filtering and mixing to generate the multi-channel television signal,wherein the RF processing block comprises a mixing stage having anoscillator with a coarse step size or a fine step size, and the digitalprocessing block is further configured to compensate for the step sizeof the oscillator when generating the known frequency shift.
 4. Thetelevision receiver of claim 1, wherein the analog processing blockcomprises at least one coarse bandpass filter or at least one coarse lowpass filter for providing analog filtering.
 5. The television receiverof claim 4, wherein the mixing stage comprises elements for providing anadditional level of analog filtering.
 6. The television receiver ofclaim 3, wherein the digital processing block is further configured toprovide equalization for compensating for non-ideal analog filtering inat least one of the analog and RF processing blocks.
 7. The televisionreceiver of claim 2, wherein the first signal is a coarse channelsignal, the second signal is a digitized coarse channel signal, and thedigital processing block is further configured to process the digitizedcoarse channel signal to provide a processed digitized coarse channelsignal, and the digital processing block comprises a video processingblock having a frequency tracking loop for tracking the carrierfrequency of the desired television channel signal, wherein thefrequency tracking loop comprises: a first frequency rotator forshifting the frequency content of the processed digitized coarse channelsignal to the baseband based on the known frequency shift andadditionally the frequency shift feedback signal for analog broadcasttransmission standards; a video filter stage for filtering the output ofthe first frequency rotator; and a digital variable gain amplifier foramplifying the output of the video filter stage.
 8. The receiver ofclaim 7, wherein the receiver further comprises a digital demodulatorcoupled to the digital processing block for processing the output of thedigital variable gain amplifier and tracking the carrier frequency ofthe desired television channel signal to provide a digital transportstream in accordance with a digital television broadcast standard. 9.The receiver of claim 8, wherein the digital demodulator is furtherconfigured to generate and provide the frequency shift feedback signalto the first frequency rotator when the desired television channelsignal is broadcast according to a digital television broadcaststandard.
 10. The receiver of claim 7, wherein the frequency trackingloop further comprises: a second frequency rotator coupled to thedigital variable gain amplifier for frequency shifting the output of thedigital variable gain amplifier; and a picture carrier recovery blockcoupled to the first and second frequency rotators, the picture carrierrecovery block being configured to receive the output of the secondfrequency rotator, generate and provide the frequency shift feedbacksignal to the first frequency rotator and provide the video informationof the desired television channel signal for analog broadcasttransmission standards.
 11. The receiver of claim 10, wherein thepicture carrier recovery block comprises: a carrier recovery filterconfigured to filter the output of the second frequency rotator toproduce a filtered picture carrier signal; a first phase rotatorconfigured to apply phase adjustment to the filtered picture carriersignal to produce a phase-adjusted filtered picture carrier signal; anda carrier recovery block coupled to the first phase rotator and thefirst frequency rotator, the carrier recovery block being configured toprocess the phase-adjusted filtered picture carrier signal to compensatefor phase noise and produce a phase control signal that is provided tothe first phase rotator to control the amount of the phase adjustment,the carrier recovery block further being configured to generate thefrequency shift feedback signal.
 12. The receiver of claim 10, whereinthe picture carrier recovery-block comprises a carrier recovery blockincluding: a phase-frequency detector configured to receive an inputphase signal based on the output of the second frequency rotator,generate a phase error signal by comparing the input phase signal with areference phase signal, and generate a frequency error signal based onthe phase error signal; a summer configured to sum the frequency errorsignal and a version of the phase error signal to produce an adjustedfrequency error signal; a frequency loop amplifier configured to providean amplified frequency error signal based on the adjusted frequencyerror signal; and a frequency oscillator block configured to generatethe frequency shift feedback signal based on the amplified frequencyerror signal.
 13. The receiver of claim 12, wherein the frequencyoscillator block comprises: a frequency accumulator configured to updatea current frequency based on the amplified frequency error signal toproduce a frequency adjusted signal; and a frequency clipping blockconfigured to specify upper and lower limits for the frequency adjustedsignal, and when the frequency adjusted signal exceeds one of thelimits, the frequency clipping block is configured to limit thefrequency adjusted signal to that limit, wherein, the frequency shiftfeedback signal is generated based on the output of the frequencyclipping block.
 14. The receiver of claim 11, wherein the carrierrecovery block further comprises: a low pass filter configured to filterthe phase error signal to produce a filtered phase error signal; and aphase loop amplifier configured to amplify the filtered phase errorsignal to produce the version of the phase error signal.
 15. Thereceiver of claim 7, wherein the digital processing block furthercomprises an audio filtering block having a second frequency trackingloop configured to extract an audio carrier frequency of the desiredtelevision channel signal for analog television broadcast standards andthe audio filtering block is configured to apply a second knownfrequency shift to compensate for a known frequency offset in the audiocarrier frequency.
 16. The receiver of claim 15, wherein the secondfrequency tracking loop comprises: a third frequency rotator forshifting the frequency content of the processed digitized coarse channelsignal to the baseband; an audio filter stage for filtering the outputof the third frequency rotator; a frequency demodulator coupled to theaudio filter stage for demodulating the output of the audio filter stageand producing a first intermediate audio signal; and an audio IF carrierrecovery block that is configured to track the audio carrier signal thatcorresponds to the desired television channel signal based on one of thefirst intermediate audio signal and a sound IF carrier recovery signal,and to produce an audio frequency shift feedback signal which isprovided to the third frequency rotator.
 17. The receiver of claim 11,wherein the digital processing block further comprises an audiofiltering block that is configured to process the output of the firstfrequency rotator to extract an audio carrier frequency of the desiredtelevision channel signal for analog television broadcast standards,wherein the processing is based on the phase control signal.
 18. Thereceiver of claim 11, wherein the digital processing block furthercomprises an audio filtering block that is configured to process theoutput of the first frequency rotator to extract an audio carrierfrequency of the desired television channel signal for analog televisionbroadcast standards, wherein the audio filtering block comprises: athird frequency rotator for shifting the frequency content of thefrequency shift feedback signal to the baseband; an audio filter stagefor filtering the output of the third frequency rotator; a frequencydemodulator for demodulating the output of the audio filter stage andproducing a first intermediate audio signal; and an audio IF carrierrecovery block that is configured to track the audio carrier signal thatcorresponds to the desired television channel signal.
 19. A method ofprocessing received television signals in a television receiver toprovide video and audio information for a desired television channelsignal, wherein the method comprises: providing coarse filtering andamplification to a multi-channel television signal to produce a firstsignal, the coarse filtering being configured to use pass bands that arewide enough to accommodate frequency shifts in the desired televisionchannel signal and analog circuitry variability in the televisionreceiver; digitizing the first signal to produce a second signal; andprocessing the second signal to obtain the video and audio informationfor the desired television channel signal by tracking a carrierfrequency of the desired television channel signal and generating andapplying a frequency shift feedback signal to compensate for frequencyshifts in the carrier frequency.
 20. The method of claim 19, wherein themethod further comprises additionally applying a known frequency shiftfor known frequency offset errors in the carrier frequency.
 21. Themethod of claim 20, wherein the method comprises utilizing an RFprocessing block for receiving broadcast television signals andproviding amplification, filtering and mixing to generate themulti-channel television signal, the RF processing block comprising amixing stage having an oscillator with a coarse step size or a fine stepsize, and the method further comprises compensating for the step size ofthe oscillator when generating the known frequency shift.
 22. The methodof claim 19, wherein the step of providing coarse filtering andamplification comprises utilizing at least one coarse bandpass filter orat least one coarse low pass filter for providing analog filtering. 23.The method of claim 22, wherein the method further comprises providingthe mixing stage with elements for providing an additional level ofanalog filtering.
 24. The method of claim 19, wherein the processingstep comprises providing equalization for compensating for non-idealanalog filtering.
 25. The method of claim 20, wherein the first signalis a coarse channel signal, the second signal is a digitized coarsechannel signal, and the method further comprises: processing thedigitized coarse channel signal to provide a processed digitized coarsechannel signal; shifting the frequency content of the processeddigitized coarse channel signal to the baseband based on the knownfrequency shift and additionally the frequency shift feedback signal foranalog broadcast transmission standards; filtering the shifted frequencycontent of the processed digitized coarse channel signal with a videofilter stage to produce a filtered signal; and amplifying the filteredsignal to produce an amplified signal.
 26. The method of claim 25,wherein the method further comprises using a digital demodulator toprocess the amplified signal and track the carrier frequency of thedesired television channel signal to provide a digital transport streamin accordance with a digital television broadcast standard.
 27. Themethod of claim 26, wherein the method further comprises using thedigital demodulator to generate and provide the frequency shift feedbacksignal when the desired television channel signal is broadcast accordingto a digital television broadcast standard.
 28. The method of claim 25,wherein for analog broadcast transmission standards, the method furthercomprises: frequency shifting the amplified signal to produce afrequency shifted amplified signal; generating the frequency shiftfeedback signal based on the frequency shifted amplified signal andproviding the frequency shift feedback signal to the first frequencyrotator; and filtering the frequency shifted amplified signal to providethe video information of the desired television channel signal.
 29. Themethod of claim 28, wherein the method further comprises: filtering thefrequency shifted amplified signal to produce a filtered picture carriersignal; applying phase adjustment to the filtered picture carrier signalto produce a phase-adjusted filtered picture carrier signal; andprocessing the phase-adjusted filtered picture carrier signal tocompensate for phase noise, to produce a phase control signal that isused to control the amount of the phase adjustment, and to generate thefrequency shift feedback signal.
 30. The method of claim 28, wherein themethod further comprises: receiving an input phase signal based on thefrequency shifted amplified signal; generating a phase error signal bycomparing the input phase signal with a reference phase signal;generating a frequency error signal based on the phase error signal;summing the frequency error signal and a version of the phase errorsignal to produce an adjusted frequency error signal; amplifying thephase adjusted frequency error signal to produce an amplified frequencyerror signal; and generating the frequency shift feedback signal basedon the amplified frequency error signal.
 31. The method of claim 30,wherein generating the frequency shift feedback signal comprises:updating a current frequency based on the amplified frequency errorsignal to produce a frequency adjusted signal; limiting the frequencyadjusted signal to an upper or lower limit when the frequency adjustedsignal exceeds the upper or lower limit respectively; and generating thefrequency shift feedback signal based on the limited frequency adjustedsignal.
 32. The method of claim 29, wherein the method furthercomprises: filtering the phase error signal to produce a filtered phaseerror signal; and amplifying the filtered phase error signal to producethe version of the phase error signal.
 33. The method of claim 25,wherein the method further comprises using a second frequency trackingloop to extract an audio carrier frequency of the desired televisionchannel signal for analog television broadcast standards by applying asecond known frequency shift to compensate for a known frequency offsetin the audio carrier frequency.
 34. The method of claim 33, wherein themethod further comprises: frequency shifting the frequency content ofthe processed digitized coarse channel signal to the baseband; filteringthe frequency shifted signal using an audio filtering stage to produce afiltered signal; demodulating the filtered signal to produce a firstintermediate audio signal; and tracking the audio carrier signal thatcorresponds to the desired television channel signal based on one of thefirst intermediate audio signal and a sound IF carrier recovery signaland producing an audio frequency shift feedback signal used in thefrequency shifting step.
 35. The method of claim 28, wherein the methodfurther comprises processing the shifted frequency content of theprocessed digitized coarse channel signal to extract an audio carrierfrequency of the desired television channel signal for analog televisionbroadcast standards, wherein the processing is based on the phasecontrol signal.
 36. The method of claim 28, wherein the method furthercomprises: frequency shifting the shifted frequency content of theprocessed digitized coarse channel signal to the baseband; filtering thefrequency shifted signal using an audio filtering stage to produce afiltered signal; demodulating the filtered signal to produce a firstintermediate audio signal; and tracking the audio carrier signal thatcorresponds to the desired television channel signal based on the firstintermediate audio signal.